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1 lt1111 33 f sumida cd54-220m 22 m h lt1111 ?ta01 m mbrs120t3 5v 100ma gnd sw2 sw1 lim i in v lt1111cs8-5 sense 3v input 10 f* m + *optional + micropower dc/dc converter adjustable and fixed 5v, 12v all surface mount 3v to 5v step-up converter d u escriptio s f ea t u re u s a o pp l ic at i typical load regulation 0 load current (ma) 0 1 output voltage (v) 2 3 4 5 6 50 100 150 200 lt1111 ?ta02 25 75 125 175 v in = 2v 2.2 2.4 2.6 2.8 3v n operates at supply voltages from 2v to 30v n 72khz oscillator n works with surface mount inductors n only three external components required n step-up or step-down mode n low-battery detector comparator on-chip n user adjustable current limit n internal 1a power switch n fixed or adjustable output voltage versions n space saving 8-pin minidip or so-8 package n 3v to 5v, 5v to 12v converters n 9v to 5v, 12v to 5v converters n remote controls n peripherals and add-on cards n battery backup supplies n uninterruptible supplies n laptop and palmtop computers n cellular telephones n portable instruments n flash memory vpp generators u a o pp l ic at i ty p i ca l the lt1111 is a versatile micropower dc/dc converter. the device requires only three external components to deliver a fixed output of 5v or 12v. supply voltage ranges from 2v to 12v in step-up mode and to 30v in step-down mode. the lt1111 functions equally well in step-up, step- down, or inverting applications. the lt1111 oscillator is set at 72khz, optimizing the device to work with off-the-shelf surface mount inductors. the device can deliver 5v at 100ma from a 3v input in step-up mode or 5v at 200ma from a 12v input in step- down mode. switch current limit can be programmed with a single resistor. an auxiliary open-collector gain block can be configured as a low-battery detector, linear post regulator, undervoltage lock-out circuit, or error amplifier. for input sources of less than 2v use the lt1110.
2 l t 1111 a u g w a w u w a r b s o lu t exi t i s operating temperature range lt1111c ............................................... 0 c to 70 c lt1111i ......................................... C 40 c to 105 c lt1111m ....................................... C 55 c to 125 c storage temperature range ................ C 65 c to 150 c lead temperature (soldering, 10 sec)................. 300 c wu u package / o rder i for atio supply voltage (v in ) ............................................... 36v sw1 pin voltage (v sw1 ) ......................................... 50v sw2 pin voltage (v sw2 ) ............................ C 0.5v to v in feedback pin voltage (lt1111) ............................. 5.5v switch current ....................................................... 1.5a maximum power dissipation ............................ 500mw t jmax = 150 c, q ja = 120 c/w (j) t jmax = 90 c, q ja = 130 c/w (n) order part number lt1111cn8 lt1111cn8-5 lt1111cn8-12 lt1111mj8 lt1111mj8-5 LT1111MJ8-12 1 2 3 4 8 7 6 5 top view i lim v in sw1 sw2 fb (sense)* set a0 gnd n8 package 8-lead plastic dip *fixed versions j8 package 8-lead ceramic dip lt1111cs8 lt1111cs8-5 lt1111cs8-12 t jmax = 90 c, q ja = 150 c/w order part number 1111 11115 11111 s8 part marking symbol parameter conditions min typ max units i q quiescent current switch off 300 400 m a v in input voltage step-up mode l 2.0 12.6 v step-down mode l 30.0 v comparator trip point voltage lt1111 (note 1) l 1.20 1.25 1.30 v v out output sense voltage lt1111-5 (note 2) l 4.75 5.00 5.25 v lt1111-12 (note 2) l 11.40 12.00 12.60 v comparator hysteresis lt1111 l 8 12.5 mv output hysteresis lt1111-5 l 32 50 mv lt1111-12 l 75 120 mv f osc oscillator frequency 54 72 88 khz dc duty cycle: step-up mode full load 43 50 59 % step-down mode 24 34 50 % t on switch on time: step-up mode i lim tied to v in 579 m s step-down mode v out , = 5v, v in = 12v 3.3 5 7.8 m s v sat sw saturation voltage, step-up mode v in = 3.0v, i sw = 650ma 0.5 0.65 v v in = 5.0v, i sw = 1a 0.8 1.0 v sw saturation voltage, step-down mode v in = 12v, i sw = 650ma 1.1 1.5 v i fb feedback pin bias current lt1111, v fb = 0v l 70 120 na i set set pin bias current v set = v ref l 70 300 na v ol gain block output low i sink = 300 m a, v set = 1.00v l 0.15 0.4 v e lectr ic al c c hara terist ics v in = 3v, military or commercial version consult factory for industrial grade parts 1 2 3 4 8 7 6 5 top view fb (sense)* set a0 gnd i lim v in sw1 sw2 s8 package 8-lead plastic so *fixed version
3 lt1111 e lectr ic al c c hara terist ics v in = 3v, military or commercial version symbol parameter conditions min typ max units reference line regulation 5v v in 30v l 0.02 0.075 %/v 2v v in 5v 0.20 0.400 %/v a v gain block gain r l = 100k (note 3) l 1000 6000 v/v i lim current limit 220 w from i lim to v in 400 ma current limit temperature coefficient l C 0.3 %/ c switch off leakage current measured at sw1 pin, v sw1 = 12v 1 10 m a maximum excursion below gnd i sw1 10 m a, switch off C 400 C 350 mv lt1111m symbol parameter conditions min typ max units i q quiescent current switch off l 300 500 m a f osc oscillator frequency l 45 72 100 khz dc duty cycle: step-up mode full load l 40 50 62 % step-down mode l 20 55 % t on switch on time: step-up mode i lim tied to v in l 5711 m s step-down mode v out = 5v, v in = 12v l 39 m s reference line regulation 2v v in 5v, 25 c t a 125 c 0.2 0.4 %/v 2.4v v in 5v, t a = C 55 c 0.8 %/v v sat sw saturation voltage, step-up mode 0 c t a 125 c, i sw = 500ma, 0.5 0.65 v t a = C 55 c, i sw = 400ma sw saturation voltage, step-down mode v in = 12v, 0 c t a 125 c 1.5 v i sw = 500ma t a = C 55 c 2.0 v v in = 3v, C 55 c t a 125 c unless otherwise noted. lt1111c symbol parameter conditions min typ max units i q quiescent current switch off l 300 450 m a f osc oscillator frequency l 54 72 95 kh dc duty cycle: step-up mode full load l 43 50 59 % step-down mode l 24 34 50 % t on switch on time: step-up mode i lim tied to v in l 5.0 7 9.0 m s step-down mode v out = 5v, v in = 12v l 3.3 5 7.8 m s reference line regulation 2v v in 5v l 0.2 0.7 %/v v sat sw saturation voltage, step-up mode v in = 3v, i sw = 650ma l 0.5 0.65 v sw saturation voltage, step-down mode v in = 12v, i sw = 650ma l 1.1 1.50 v v in = 3v, 0 c t a 70 c unless otherwise noted. the l denotes specifications which apply over the full operating temperature range. note 1: this specification guarantees that both the high and low trip points of the comparator fall within the 1.20v to 1.30v range. note 2: the output voltage waveform will exhibit a sawtooth shape due to the comparator hysteresis. the output voltage on the fixed output versions will always be within the specified range. note 3: 100k resistor connected between a 5v source and the a0 pin.
4 l t 1111 cc hara terist ics uw a t y p i ca lper f o r c e temperature ( c) on time ( m s) 50 ?5 0 25 lt111 ?tpc03 50 75 100 125 10 9.5 9.0 8.5 8.0 7.5 7.0 6.5 6.0 5.5 5.0 oscillator frequency oscillator frequency switch on time temperature ( c) ?0 40 oscillator frequency (khz) 50 60 70 80 90 100 ?5 0 25 50 lt1111 ?tpc01 75 100 125 input voltage (v) 0 69 frequency (khz) 70 71 72 73 74 75 36 912 lt1111 ?tpc02 15 18 21 68 67 24 27 30 saturation voltage saturation voltage duty cycle step-up mode step-up mode temperature ( c) duty cycle (%) 50 ?5 0 25 lt1111 ?tpc04 50 75 100 125 60 58 56 54 52 50 48 46 44 42 40 switch current (a) saturation voltage (v) 0 0.2 0.4 0.6 lt1111 ?tpc06 0.8 1.0 1.2 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 1.4 v in = 3v v in = 2v v in = 5v temperature ( c) saturation voltage (v) 50 25 0 25 lt1111 ?tpc05 50 75 100 125 1.0 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 v in = 3v i sw = 650ma switch on voltage switch on voltage minimum/maximum frequency step-down mode step-down mode vs on time switch current (a) on voltage (v) 0 0.2 0.4 0.6 lt1111 ?tpc08 0.8 1.0 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 v in = 12v temperature ( c) on voltage (v) 50 ?5 0 25 lt1111 ?tpc07 50 75 100 2.00 1.75 1.50 1.25 1.00 0.75 0.50 125 v in = 12v i sw = 650ma switch on time ( m s) oscillator frequency (khz) 4 567 lt1111 ?tpc09 89 100 90 80 70 60 50 40 10 11 12 ?5 c t a 125 c 0 c t a 70 c
5 lt1111 cc hara terist ics uw a t y p i ca lper f o r c e gnd (pin 5): ground. a0 (pin 6): auxiliary gain block (gb) output. open collector, can sink 300 m a. set (pin 7): gb input. gb is an op amp with positive input connected to set pin and negative input connected to 1.25v reference. fb/sense (pin 8): on the lt1111 (adjustable) this pin goes to the comparator input. on the lt1111-5 and lt1111-12, this pin goes to the internal application resistor that sets output voltage. maximum switch current quiescent current quiescent current vs r lim input voltage (v) quiescent current ( m a) 0 lt1111 ?tpc10 3 400 380 360 340 320 280 260 6 240 220 200 300 9 12151821242730 temperature ( c) quiescent current ( m a) ?0 lt1111 ?tpc11 500 450 400 350 250 ?5 150 100 200 300 0 25 50 75 100 125 temperature ( c) bias current (na) ?0 lt1111 ?tpc14 100 90 80 70 50 ?5 30 20 40 60 0 25 50 75 100 125 10 0 i lim (pin 1): connect this pin to v in for normal use. where lower current limit is desired, connect a resistor between i lim and v in . a 220 w resistor will limit the switch current to approximately 400ma. v in (pin 2): input supply voltage. sw1 (pin 3): collector of power transistor. for step-up mode connect to inductor/diode. for step-down mode connect to v in . sw2 (pin 4): emitter of power transistor. for step-up mode connect to ground. for step-down mode connect to inductor/diode. this pin must never be allowed to go more than a schottky diode drop below ground. pi u fu u c u s o ti temperature ( c) bias current (na) ?0 lt1111 ?tpc13 100 90 80 70 50 ?5 30 20 40 60 0 25 50 75 100 125 10 0 set pin bias current feedback bias current r lim ( w ) switch current (a) 10 lt1111 ?tpc12 100 1.5 1.4 1.3 1.2 1.1 0.9 0.8 1000 0.7 0.6 0.5 0.4 0.3 0.2 0.1 1.0 step-down v in = 12v step-up 2v v in 5v
6 l t 1111 gain block a2 can serve as a low-battery detector. the negative input of a2 is the 1.25v reference. a resistor divider from v in to gnd, with the mid-point connected to the set pin provides the trip voltage in a low-battery detector application. ao can sink 300 m a (use a 22k resistor pull-up to 5v). a resistor connected between the i lim pin and v in sets maximum switch current. when the switch current ex- ceeds the set value, the switch cycle is prematurely terminated. if current limit is not used, i lim should be tied directly to v in . propagation delay through the current limit circuitry is approximately 1 m s. in step-up mode the switch emitter (sw2) is connected to ground and the switch collector (sw1) drives the induc- tor; in step-down mode the collector is connected to v in and the emitter drives the inductor. the lt1111-5 and lt1111-12 are functionally identical to the lt1111. the -5 and -12 versions have on-chip voltage setting resistors for fixed 5v or 12v outputs. pin 8 on the fixed versions should be connected to the output. no external resistors are needed. u lt11 o at i 11 oper the lt1111 is a gated oscillator switcher. this type architecture has very low supply current because the switch is cycled when the feedback pin voltage drops below the reference voltage. circuit operation can best be understood by referring to the lt1111 block diagram. comparator a1 compares the feedback (fb) pin voltage with the 1.25v reference signal. when fb drops below 1.25v, a1 switches on the 72khz oscillator. the driver amplifier boosts the signal level to drive the output npn power switch. the switch cycling action raises the output voltage and fb pin voltage. when the fb voltage is suffi- cient to trip a1, the oscillator is gated off. a small amount of hysteresis built into a1 ensures loop stability without external frequency compensation. when the comparator output is low, the oscillator and all high current circuitry is turned off, lowering device quiescent current to just 300 m a. the oscillator is set internally for 7 m s on time and 7 m s off time, optimizing the device for circuits where v out and v in differ by roughly a factor of 2. examples include a 3v to 5v step-up converter or a 9v to 5v step-down converter. w i dagra b l o c k s lt1111 ?bd01 in v gnd set a0 a2 1.25v reference a1 oscillator driver sw1 sw2 lim i gain block/ error amp comparator + + fb lt1111-5/lt1111-12 lt1111 lt1111 ?bd02 in v gnd set a0 a2 1.25v reference a1 oscillator driver r1 sw1 sw2 lim i r2 220k sense lt1111-5: lt1111-12: r1 = 73.5k r1 = 25.5k gain block/ error amp comparator + +
7 lt1111 inductor selection general a dc/dc converter operates by storing energy as mag- netic flux in an inductor core, and then switching this energy into the load. since it is flux, not charge, that is stored, the output voltage can be higher, lower, or oppo- site in polarity to the input voltage by choosing an appropriate switching topology. to operate as an efficient energy transfer element, the inductor must fulfill three requirements. first, the inductance must be low enough for the inductor to store adequate energy under the worst case condition of minimum input voltage and switch-on time. the inductance must also be high enough so maxi- mum current ratings of the lt1111 and inductor are not exceeded at the other worst case condition of maximum input voltage and on time. additionally, the inductor core must be able to store the required flux; i.e., it must not saturate. at power levels generally encountered with lt1111 based designs, small surface mount ferrite core units with saturation current ratings in the 300ma to 1a range and dcr less than 0.4 w (depending on application) are adequate. lastly, the inductor must have sufficiently low dc resistance so excessive power is not lost as heat in the windings. an additional consideration is electro- magnetic interference (emi). toroid and pot core type inductors are recommended in applications where emi must be kept to a minimum; for example, where there are sensitive analog circuitry or transducers nearby. rod core types are a less expensive choice where emi is not a problem. minimum and maximum input voltage, output voltage and output current must be established before an inductor can be selected. inductor selection step-up converter in a step-up, or boost converter (figure 4), power gener- ated by the inductor makes up the difference between input and output. power required from the inductor is determined by: pv vv i l out d in min out =+ ()() () 01 where v d is the diode drop (0.5v for a 1n5818 schottky). energy required by the inductor per cycle must be equal or greater than: u s a o pp l ic at i wu u i for atio p f l osc /() 02 in order for the converter to regulate the output. when the switch is closed, current in the inductor builds according to: it v r e l in rt l () ( ) = ? ? ? ? 103 where r is the sum of the switch equivalent resistance (0.8 w typical at 25 c) and the inductor dc resistance. when the drop across the switch is small compared to v in , the simple lossless equation: it v l t l in () = () 04 can be used. these equations assume that at t = 0, inductor current is zero. this situation is called discon- tinuous mode operation in switching regulator parlance. setting t to the switch-on time from the lt1111 speci- fication table (typically 7 m s) will yield i peak for a specific l and v in . once i peak is known, energy in the inductor at the end of the switch-on time can be calculated as: eli l peak = 1 2 05 2 () e l must be greater than p l /f osc for the converter to deliver the required power. for best efficiency i peak should be kept to 1a or less. higher switch currents will cause excessive drop across the switch resulting in reduced efficiency. in general, switch current should be held to as low a value as possible in order to keep switch, diode and inductor losses at a minimum. as an example, suppose 12v at 60ma is to be generated from a 4.5v to 8v input. recalling equation (01), p v v v ma mw l =+ ()() = 12 0 5 4 5 60 480 06 .. () energy required from the inductor is p f mw khz j l osc == 480 72 67 07 .() m
8 l t 1111 u s a o pp l ic at i wu u i for atio i out = output current v out = output voltage v in = minimum input voltage v sw is actually a function of switch current which is in turn a function of v in , l, time, and v out . to simplify, 1.5v can be used for v sw as a very conservative value. once i peak is known, inductor value can be derived from: l vvv i t in min sw out peak on = -- () 11 where t on = switch-on time (7 m s). next, the current limit resistor r lim is selected to give i peak from the r lim step-down mode curve. the addition of this resistor keeps maximum switch current constant as the input voltage is increased. as an example, suppose 5v at 300ma is to be generated from a 12v to 24v input. recalling equation (10), i ma ma peak = () + + ? ? = 2 300 050 505 12 15 05 600 12 . . . . () next, inductor value is calculated using equation (11): l ma sh == 12 1 5 5 600 764 13 . .() mm use the next lowest standard value (56 m h). then pick r lim from the curve. for i peak = 600ma, r lim = 56 w . inductor selection positive-to-negative converter figure 7 shows hookup for positive-to-negative conver- sion. all of the output power must come from the inductor. in this case, p l = ( ? v out ? + v d )( i out ) (14) in this mode the switch is arranged in common collector or step-down mode. the switch drop can be modeled as a 0.75v source in series with a 0.65 w resistor. when the picking an inductor value of 47 m h with 0.2 w dcr results in a peak switch current of: i v ema peak s h = ? ? ? ? = 45 10 1 623 08 10 7 47 . . .() . w wm m substituting i peak into equation 04 results in: ehaj l = ()( ) = 1 2 47 0 623 9 1 09 2 mm .. () since 9.1 m j > 6.7 m j, the 47 m h inductor will work. this trial-and-error approach can be used to select the opti- mum inductor. keep in mind the switch current maximum rating of 1.5a. if the calculated peak current exceeds this, consider using the lt1110. the 70% duty cycle of the lt1110 allows more energy per cycle to be stored in the inductor, resulting in more output power. a resistor can be added in series with the i lim pin to invoke switch current limit. the resistor should be picked so the calculated i peak at minimum v in is equal to the maximum switch current (from typical performance characteristic curves). then, as v in increases, switch current is held constant, resulting in increasing efficiency. inductor selection step-down converter the step-down case (figure 5) differs from the step-up in that the inductor current flows through the load during both the charge and discharge periods of the inductor. current through the switch should be limited to ~650ma in this mode. higher current can be obtained by using an external switch (see figure 6). the i lim pin is the key to successful operation over varying inputs. after establishing output voltage, output current and input voltage range, peak switch current can be calculated by the formula: i i dc vv vv v peak out out d in sw d = + + ? ? 2 10 () where dc = duty cycle (0.50) v sw = switch drop in step-down mode v d = diode drop (0.5v for a 1n5818)
9 lt1111 capacitors provide still better performance at more ex- pense. we recommend os-con capacitors from sanyo corporation (san diego, ca). these units are physically quite small and have extremely low esr. to illustrate, figures 1, 2, and 3 show the output voltage of an lt1111 based converter with three 100 m f capacitors. the peak switch current is 500ma in all cases. figure 1 shows a sprague 501d, 25v aluminum capacitor. v out jumps by over 120mv when the switch turns off, followed by a drop in voltage as the inductor dumps into the capacitor. this works out to be an esr of over 0.24 w . figure 2 shows the same circuit, but with a sprague 150d, 20v tantalum capacitor replacing the aluminum unit. output jump is now about 35mv, corresponding to an esr of 0.07 w . figure 3 shows the circuit with a 16v os-con unit. esr is now only 0.02 w . u s a o pp l ic at i wu u i for atio switch closes, current in the inductor builds according to it v r e l l rt l () = ? ? ? ? 115 () where r = 0.65 w + dcr l v l = v in C 0.75v as an example, suppose C5v at 50ma is to be generated from a 4.5v to 5.5v input. recalling equation (14), p l = ( ? -5v ? +0.5v )( 50ma ) = 275mw (16) energy required from the inductor is: p f mw khz j l osc == 275 72 38 17 .. () m picking an inductor value of 56 m h with 0.2 w dcr results in a peak switch current of: i vv ema peak s h = () + () ? ? ? ? = 45 075 065 02 1 445 18 085 7 56 .. .. .() . ww wm m substituting i peak into equation (04) results in: ehaj l = ()( ) = 1 2 56 0 445 5 54 19 2 mm ...() since 5.54 m j > 3.82 m j, the 56 m h inductor will work. with this relatively small input range, r lim is not usually necessary and the i lim pin can be tied directly to v in . as in the step-down case, peak switch current should be limited to ~650ma. capacitor selection selecting the right output capacitor is almost as important as selecting the right inductor. a poor choice for a filter capacitor can result in poor efficiency and/or high output ripple. ordinary aluminum electrolytics, while inexpensive and readily available, may have unacceptably poor equiva- lent series resistance (esr) and esl (inductance). there are low esr aluminum capacitors on the market specifi- cally designed for switch mode dc/dc converters which work much better than general-purpose units. tantalum 50mv/div 5 m s/div lt1111 ? f01 figure 1. aluminum 50mv/div 5 m s/div lt1111 ? f02 figure 2. tantalum 50mv/div 5 m s/div lt1111 ? f01 figure 3. os-con
10 l t 1111 u s a o pp l ic at i wu u i for atio diode selection speed, forward drop, and leakage current are the three main considerations in selecting a catch diode for lt1111 converters. general purpose rectifiers such as the 1n4001 are unsuitable for use in any switching regulator applica- tion. although they are rated at 1a, the switching time of a 1n4001 is in the 10 m s to 50 m s range. at best, efficiency will be severely compromised when these diodes are used; at worst, the circuit may not work at all. most lt1111 circuits will be well served by a 1n5818 schottky diode, or its surface mount equivalent, the mbrs130t3. the combination of 500mv forward drop at 1a current, fast turn on and turn off time, and 4 m a to 10 m a leakage current fit nicely with lt1111 requirements. at peak switch currents of 100ma or less, a 1n4148 signal diode may be used. this diode has leakage current in the 1na to 5na range at 25 c and lower cost than a 1n5818. (you can also use them to get your circuit up and running, but beware of destroying the diode at 1a switch currents.) step-up (boost mode) operation a step-up dc/dc converter delivers an output voltage higher than the input voltage. step-up converters are not short-circuit protected since there is a dc path from input to output. the usual step-up configuration for the lt1111 is shown in figure 4. the lt1111 first pulls sw1 low causing v in C v cesat to appear across l1. a current then builds up in l1. at the end of the switch on time the current in l1 is 1 : i v l t pea k in on = () 20 immediately after switch turn-off, the sw1 voltage pin starts to rise because current cannot instantaneously stop flowing in l1. when the voltage reaches v out + v d , the inductor current flows through d1 into c1, increasing v out . this action is repeated as needed by the lt1111 to keep v fb at the internal reference voltage of 1.25v. r1 and r2 set the output voltage according to the formula v r r v out =+ ? ? ? ? () 1 2 1 125 21 .() step-down (buck mode) operation a step-down dc/dc converter converts a higher voltage to a lower voltage. the usual hookup for an lt1111 based step-down converter is shown in figure 5. l1 lt1111 ?f04 gnd sw2 sw1 lim i in v d1 r3* lt1111 + v out r2 r1 c1 *optional v in fb figure 4. step-up mode hookup. refer to table 1 for component values. note 1: this simple expression neglects the effect of switch and coil resistance. this is taken into account in the inductor selection section. when the switch turns on, sw2 pulls up to v in C v sw . this puts a voltage across l1 equal to v in C v sw C v out , causing a current to build up in l1. at the end of the switch on time, the current in l1 is equal to: i v vv l t peak in sw out on = -- () 22 lt1111 ?f05 gnd sw2 sw1 lim i in v r3 100 fb v out + c2 + c1 d1 1n5818 v in r2 r1 l1 w lt1111 figure 5. step-down mode hookup
11 lt1111 u s a o pp l ic at i wu u i for atio figure 6. q1 permits higher current switching. lt1111 functions as controller. when the switch turns off, the sw2 pin falls rapidly and actually goes below ground. d1 turns on when sw2 reaches 0.4v below ground. d1 must be a schottky diode . the voltage at sw2 must never be allowed to go below C0.5v. a silicon diode such as the 1n4933 will allow sw2 to go to C0.8v, causing potentially destructive power dissipation inside the lt1111. output voltage is deter- mined by: v r r v out =+ ? ? ? ? () 1 2 1 125 23 .() r3 programs switch current limit. this is especially impor- tant in applications where the input varies over a wide range. without r3, the switch stays on for a fixed time each cycle. under certain conditions the current in l1 can build up to excessive levels, exceeding the switch rating and/or saturating the inductor. the 100 w resistor programs the switch to turn off when the current reaches approximately 700ma. when using the lt1111 in step-down mode, output voltage should be limited to 6.2v or less. higher output voltages can be accommodated by inserting a 1n5818 diode in series with the sw2 pin (anode con- nected to sw2). higher current step-down operation output current can be increased by using a discrete pnp pass transistor as shown in figure 6. r1 serves as a current limit sense. when the voltage drop across r1 equals a v be , the switch turns off. for temperature com- pensation a schottky diode can be inserted in series with the i lim pin. this also lowers the maximum drop across r1 to v be C v d , increasing efficiency. as shown, switch current is limited to 2a. inductor value can be calculated based on formulas in the inductor selection step- down converter section with the following conservative expression for v sw : vvv v sw r q sat =+ ? 11 10 24 .() r2 provides a current path to turn off q1. r3 provides base drive to q1. r4 and r5 set output voltage. a pmos fet can be used in place of q1 when v in is between 10v and 20v. lt1111 ?ta08 d1 1n5821 + + v out v in 30v max l1 r1 0.3 w r2 220 q1 mje210 or zetex ztx749 r3 330 r4 r5 c1 v out = 1.25v ( 1 + ) r4 r5 lt1111 gnd sw2 sw1 v in i l fb c2 in figure 8, the input is negative while the output is positive. in this configuration, the magnitude of the input voltage can be higher or lower than the output voltage. a level shift, provided by the pnp transistor, supplies proper polarity feedback information to the regulator. inverting configurations the lt1111 can be configured as a positive-to-negative converter (figure 7), or a negative-to-positive converter (figure 8). in figure 7, the arrangement is very similar to a step-down, except that the high side of the feedback is referred to ground. this level shifts the output negative. as in the step-down mode, d1 must be a schottky diode, and ? v out ? should be less than 6.2v. more negative out- put voltages can be accommodated as in the prior section. lt1111 ?f07 ? out + c2 + c1 d1 1n5818 v in r1 r2 l1 gnd sw2 sw1 lim i in v r3 fb lt1111 figure 7. positive-to-negative converter
12 l t 1111 u s a o pp l ic at i wu u i for atio l1 lt1111 ?f08 gnd sw2 fb sw1 lim i in v d1 a0 v out r2 v out = 1.25v + 0.6v r1 r2 ( ) r1 2n3906 ? in + c1 lt1111 + c2 figure 8. negative-to-positive converter using the i lim pin the lt1111 switch can be programmed to turn off at a set switch current, a feature not found on competing devices. this enables the input to vary over a wide range without exceeding the maximum switch rating or saturating the inductor. consider the case wh ere analysis shows the lt1111 must operate at an 800ma peak switch current with a 2v input. if v in rises to 4v, the peak switch current will rise to 1.6a, exceeding the maximum switch current rating. with the proper resistor selected (see the maxi- mum switch current vs i lim characteristic), the switch current will be limited to 800ma, even if the input voltage increases. another situation where the i lim feature is useful occurs when the device goes into continuous mode operation. this occurs in step-up mode when: v v vv dc out diode in sw + - < - 1 1 25 () when the input and output voltages satisfy this relation- ship, inductor current does not go to zero during the switch off time. when the switch turns on again, the current ramp starts from the non-zero current level in the inductor just prior to switch turn-on. as shown in figure 9, the inductor current increases to a high level before the comparator turns off the oscillator. this high current can cause excessive output ripple and requires oversizing the output capacitor and inductor. with the i lim feature, however, the switch current turns off at a programmed level as shown in figure 10, keeping output ripple to a minimum. lt1111 ?f09 i off l on switch figure 9. no current limit causes large inductor current build-up lt1111 ?f10 i on l off switch programmed current limit figure 10. current limit keeps inductor current under control figure 11 details current limit circuitry. sense transistor q1, whose base and emitter are paralleled with power switch q2, is ratioed such that approximately 0.5% of q2s collector current flows in q1s collector. this current is passed through internal 80 w resistor r1 and out through the i lim pin. the value of the external resistor connected between i lim and v in sets the current limit. when sufficient switch current flows to develop a v be across r1 + r lim , q3 turns on and injects current into the oscillator, turning off the switch. delay through this cir- cuitry is approximately 1 m s. the current trip point be- comes less accurate for switch on times less than 3 m s. resistor values programming switch on time for 1 m s or less will cause spurious response in the switch circuitry although the device will still maintain output regulation. lt1111 ?f11 sw2 sw1 driver oscillator v in i lim r1 80 w (internal) r lim (external) q1 q2 q3 figure 11. lt1111 current limit circuitry
13 lt1111 u s a o pp l ic at i wu u i for atio table 2. inductor manufacturers manufacturer part numbers coiltronics incorporated ctx100-4 series 6000 park of commerce blvd. surface mount boca raton, fl 33487 407-241-7876 toko america incorporated type 8rbs 1250 feehanville drive mount prospect, il 60056 312-297-0070 sumida electric co. usa cd54 708-956-0666 cdr74 cdr105 surface mount using the gain block the gain block (gb) on the lt1111 can be used as an error amplifier, low-battery detector or linear post regulator. the gain block itself is a very simple pnp input op amp with an open collector npn output. the negative input of the gain block is tied internally to the 1.25v reference. the positive input comes out on the set pin. arrangement of the gain block as a low-battery detector is straightforward. figure 12 shows hookup. r1 and r2 need only be low enough in value so that the bias current of the set input does not cause large errors. 33k for r2 is adequate. r3 can be added to introduce a small amount of hysteresis. this will cause the gain block to snap when the trip point is reached. values in the 1m to 10m range are optimal. however, the addition of r3 will change the trip point. table 3. capacitor manufacturers manufacturer part numbers sanyo video components os-con series 1201 sanyo avenue san diego, ca 92073 619-661-6322 nichicon america corporation pl series 927 east state parkway schaumberg, il 60173 708-843-7500 sprague electric company 150d solid tantalums lower main street 550d tantalex sanford, me 04073 207-324-4140 matsuo 267 series 714-969-2491 surface mount lt1111 ?f12 v bat r1 r2 1.25v ref set gnd in v lt1111 47k 5v to processor + a0 r3 r1 = v lb ?1.25v 35.1 m a v lb = battery trip point r2 = 33k r3 = 1.6m figure 12. setting low-battery detector trip point table 1. component selection for common converters input output output circuit inductor inductor capacitor voltage voltage current (min) figure value part number value notes 2 to 3.1 5 90ma 4 15 m h s cd75-750k 33 m f* 2 to 3.1 5 10ma 4 47 m h s cd54-470k, c ctx50-1 10 m f 2 to 3.1 12 30ma 4 15 m h s cd75-150k 22 m f 2 to 3.1 12 10ma 4 47 m h s cd54-470k, c ctx50-1 10 m f 5 12 90ma 4 33 m h s cd75-330k 22 m f 5 12 30ma 4 47 m h s cd75-470k, c ctx50-1 15 m f 6.5 to 11 5 50ma 5 15 m h s cd54-150k 47 m f** 12 to 20 5 300ma 5 56 m h s cd105-560k, c ctx50-4 47 m f** 20 to 30 5 300ma 5 120 m h s cd105-121k, c ctx100-4 47 m f** 5 C5 75ma 6 56 m h s cd75-560k, c ctx50-4 47 m f 12 C5 250ma 6 120 m h s cd105-121k, c ctx100-4 100 m f** s = sumida c = coiltronics * add 47 w from i lim to v in ** add 220 w from i lim to v in
14 l t 1111 u s a o pp l ic at i ty p i ca l 3v to C 22v lcd bias generator 9v to 5v step-down converter 20v to 5v step-down converter in v sw2 sw1 lt1111 ?ta03 lim i gnd r1 100 w lt1111 mbrs130t3 4.7 m f l1* 27 m h + ?2v output 7ma at 2v input * l1 = sumida cd54-270k for 5v input change r1 to 47 w . converter will deliver ?2v at 40ma. 2 1.5v cells 3v fb 22 m f + 220k 0.1 m f 1n4148 39.2k 1% 732k 1% mbrs130t3 in v sw2 sw1 lt1111 ?ta04 lim i sense gnd 100 9v battery lt1111-5 mbrs130t3 22 m f l1* 15 m h w + 5v output 150ma at 9v input 50ma at 6.5v input * l1 = sumida cd54-150k in v sw2 sw1 lt1111 ?ta06 lim i sense gnd 100 lt1111-5 mbrs130t3 47 m f l1* 68 m h w + 5v output 300ma * l1 = sumida cd74-680m v in 12v to 28v
15 lt1111 u s a o pp l ic at i ty p i ca l information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 5v to C5v converter lt1111 ?ta20 lim i fb gnd lt1111 v in 8v to 18v bat54 l1* 10 m h, 3a + 220 m f 5v 500ma 0.22 w 51 w 2k 1n4148 40.2k 121k operate standby in v sw1 sw2 mtm20p08 mbrd320 * l1 = sumida cdr105-100m 2n3904 high power, low quiescent current step-down converter in v sw2 sw1 lt1111 ?ta05 lim i sense gnd 100 lt1111-5 mbrs130t3 33 m f l1* 33 m h w + 5v output 75ma 22 m f + v in 5v input * l1 = sumida cd54-330k voltage controlled positive-to-negative converter lt1111 ?ta07 fb gnd 220 w lt1111 * l1 = coiltronics ctx20-4 ? zetex inc. 516-543-7100 v in 5v to 12v bat54 220 w l1* 20 m h, 3a + in v lt1006 + 47 m f 39k 200k ? out = ?.13 v c 2w maximum output v c (0v to 5v) 0.22 w mbrd320 in v sw2 sw1 lim i zetex ? ztx788a
16 l t 1111 linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7487 (408) 432-1900 l fax : (408) 434-0507 l telex : 499-3977 ? linear technology corporation 1994 lt/gp 0594 5k rev c ? printed in usa u package d e sc r i pti o dimensions in inches (millimeters) unless otherwise noted. j8 package 8-lead ceramic dip n8 package 8-lead plastic dip 0.290 ?0.320 (7.366 ?8.128) 0.008 ?0.018 (0.203 ?0.457) 0??15 0.385 ?0.025 (9.779 ?0.635) 0.005 (0.127) min 0.405 (10.287) max 0.220 ?0.310 (5.588 ?7.874) 12 3 4 87 65 0.025 (0.635) rad typ 0.045 ?0.068 (1.143 ?1.727) full lead option 0.023 ?0.045 (0.584 ?1.143) half lead option corner leads option (4 plcs) 0.014 ?0.026 (0.360 ?0.660) 0.200 (5.080) max 0.015 ?0.060 (0.381 ?1.524) 0.125 3.175 min 0.100 ?0.010 (2.540 ?0.254) 0.045 ?0.068 (1.143 ?1.727) note: lead dimensions apply to solder dip or tin plate leads. 0.009 ?0.015 (0.229 ?0.381) 0.300 ?0.320 (7.620 ?8.128) 0.325 +0.025 0.015 +0.635 0.381 8.255 () 0.045 ?0.015 (1.143 ?0.381) 0.100 ?0.010 (2.540 ?0.254) 0.065 (1.651) typ 0.045 ?0.065 (1.143 ?1.651) 0.130 ?0.005 (3.302 ?0.127) 0.020 (0.508) min 0.018 ?0.003 (0.457 ?0.076) 0.125 (3.175) min 12 3 4 87 6 5 0.250 ?0.010 (6.350 ?0.254) 0.400 (10.160) max s8 package 8-lead plastic soic so8 0294 0.016 ?0.050 0.406 ?1.270 0.010 ?0.020 (0.254 ?0.508) 45 0 8?typ 0.008 ?0.010 (0.203 ?0.254) 0.053 ?0.069 (1.346 ?1.752) 0.014 ?0.019 (0.355 ?0.483) 0.004 ?0.010 (0.101 ?0.254) 0.050 (1.270) bsc 1 2 3 4 0.150 ?0.157 (3.810 ?3.988) 8 7 6 5 0.189 ?0.197 (4.801 ?5.004) 0.228 ?0.244 (5.791 ?6.197) *these dimensions do not include mold flash or protrusions. mold flash or protrusions shall not exceed 0.006 inch (0.15mm).

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