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linkswitch-tn design guide application note AN-37 january 2004 figure 1 (a). basic configuration using linkswitch-tn in a buck converter. figure 1 (b) basic configuration using linkswitch-t n in a buck-boost converter. introduction linkswitch-tn combines a high voltage power mosfet switch with an on/off controller in one device. it is completely self- powered from the drain pin, has a jittered switching frequency for low emi and is fully fault protected. auto-restart limits device and circuit dissipation during overload and output short circuit while over temperature protection disables the internal mosfet during thermal faults. the high thermal shutdown threshold is ideal for applications where the ambient temperature is high while the large hysteresis protects the pcb and surrounding components from high average temperatures. linkswitch-tn is designed for any application where a non- isolated supply is required such as appliances (coffee machines, rice cookers, dishwashers, microwave ovens etc.), nightlights, emergency exit signs and led drivers. linkswitch-tn can be configured in all common topologies to give a line or neutral referenced output and an inverted or non-inverted output voltage - ideal for applications using triacs for ac load control. using a switching power supply rather than a passive dropper (capacitive or resistive) gives a number of advantages, some of which are listed below. universal input ?the same power supply/product can be used worldwide high power density ?smaller size, no f? of x class capacitance needed high efficiency ?full load efficiencies >75% typical for 12 v output excellent line and load regulation high efficiency at light load ?on/off control maintains high efficiency even at light load extremely energy efficient ?input power <100 mw at no load entirely manufacturable in smd ? ore robust to drop test mechanical shock fully fault protected (overload, short circuit and thermal faults) scalable ? linkswitch-tn family allows the same basic design to be used from <50 ma to 360 ma scope this application note is for engineers designing a non-isolated power supply using the linkswitch-tn family of devices. this 1 (a) 1 (b) v o r f d in2 l in d in2 c in1 c in2 r fb d fb r bias c fb c bp d fw c o r pl ac input pi-3764-121003 + fb bp s d linkswitch-tn l v o ac input pi-3765-121003 + fb bp s d linkswitch-tn r f d in1 l in c bp r bias c fb r fb l d fw c o r pl d fb d in2 c in1 c in2
AN-37 2 a 1/04 t able 1. linkswitch-tn circuit configurations using directly sensed feedback. t opology basic circuit schematic key features high-side 1) output referenced to input buck 2) positive output (v o ) with respect to -v in direct 3) step down ?v o < v in feedback 4) low cost direct feedback ( 10% typ.) high-side 1) output referenced to input buck boost 2) negative output (v o ) with respect to -v in direct 3) step down ?v o > v in or v o < v in feedback 4) low cost direct feedback ( 10% typ.) 5) fail-safe ?output is not subjected to input voltage if the internal mosfet fails 6) ideal for driving leds ?better accuracy and temperature stability than low-side buck constant current led driver v o v in pi-3751-121003 + + fb bp s d linkswitch-tn document describes the design procedure for buck and buck- boost converters using the linkswitch-tn family of integrated off-line switchers. the objective of this document is to provide power supply engineers with guidelines in order to enable them to quickly build efficient and low cost buck or buck-boost converter based power supplies using low cost off-the-shelf inductors. complete design equations are provided for the selection of the converter? key components. since the power mosfet and controller are integrated into a single ic the design process is greatly simplified, the circuit configuration has few parts and no transformer is required. therefore a quick start section is provided that allows off-the-shelf components to be selected for common output voltages and currents. in addition to this application note a design spreadsheet is available within the pixls tool in the pi expert design software suite. the reader may also find the linkswitch-tn dak engineering prototype board useful as an example of a working supply. further details of support tools and updates to this document can be found at www.powerint.com. quick start readers wanting to start immediately can use the following information to quickly select the components for a new design, using figure 1 and tables 1 and 2 as references. 1) for ac input designs select the input stage (table 9). 2) select the topology (tables 1 and 2). - if better than 10% output regulation is required, then use opto coupler feedback with suitable reference. 3) select the linkswitch-tn device, l, r fb or v z , r bias , c fb , r z and the reverse recovery time for d fw (table 3: buck, table 4:buck-boost). 4) select freewheeling diode to meet t rr determined in step 3 (table 5). 5) for direct feedback designs, if the minimum load < 3 ma then calculate r pl = v o / 3 ma. 6) select c o as 100 f, 1.25 v o , low esr type. 7) construct prototype and verify design. notes 1. low cost, directly sensed feedback typically achieves overall regulation tolerance of 10%. 2. to ensure output regulation a pre-load may be required to maintain a minimum load current of 3 ma (buck and buck-boost only). 3. boost topology (step up) also possible but not shown. v o v in pi-3794-121503 + + fb bp s d linkswitch-tn
AN-37 3 a 1/04 t able 2. linkswitch-tn circuit configurations using optocoupler feedback. t opology basic circuit schematic key features high-side 1) output referenced to input buck 2) positive output (v o ) with respect to -v in optocoupler 3) step down ?v o < v in feedback 4) optocoupler feedback - accuracy only limited by reference choice - low cost non-safety rated optocoupler - no pre-load required 5) minimum no-load consumption low-side 1) output referenced to input buck 2) negative output (v o ) with respect to +v in optocoupler 3) step down ?v o < v in feedback 4) optocoupler feedback - accuracy only limited by reference choice - low cost non-safety rated optocoupler - no pre-load required linkswitch-tn pi-3796-121903 + + bp fb d s v o v z r z v in linkswitch-tn pi-3797-121903 + + bp fb d s v o v in v z r z low-side 1) output referenced to input buck boost 2) positive output (v o ) with respect to +v in optocoupler 3) step up/down ?v o > v in or v o < v in feedback 4) optocoupler feedback - accuracy only limited by reference choice - low cost non-safety rated optocoupler - no pre-load required 5) fail-safe ?output is not subjected to input voltage if the internal mosfet fails 6) minimum no-load consumption linkswitch-tn pi-3798-121903 + bp fb d s v o v in + v z r z notes 1. performance of opto feedback only limited by accuracy of reference (zener or ic). 2. optocoupler does not need to be safety approved. 3. reference bias current provides minimum load. the value of r z is determined by zener test current or reference ic bias current. t ypically 470 ? to 2 k ? , 1/8 w, 5% 4. boost topology (step-up) is also possible but not shown. 5. optocoupler feedback provides lowest no-load consumption.
AN-37 4 a 1/04 t able 3. components quick select for buck converters. v out i out(max) hi rms (ma) tokin coilcraft lnk30x mode diode t rr r fb v z 5 120 160 175 225 280 360 680 220 680 230 680 320 680 340 680 440 680 430 sbc2-681-211 sbc2-681-211 sbc3-681-211 sbc4-681-211 sbc4-681-211 sbc4-681-211 rfb0807-681 rfb0807-681 rfb0810-681 rfb0810-681 rfb0810-681 rfb0810-681 lnk304 lnk305 lnk306 mdcm ccm mdcm ccm mdcm ccm 75 ns 35 ns 75 ns 35 ns 75 ns 35 ns 3.84 k ? 3.9 v inductor 12 85 120 160 175 225 280 360 680 180 1000 230 1500 320 680 340 1000 440 680 430 1500 400 sbc2-681-211 sbc3-102-281 sbc3-152-251 sbc3-681-361 sbc4-102-291 sbc4-681-431 sbc6-152-451 rfb0807-681 rfb0807-102 rfb0810-152 rfb0810-681 rfb0810-102 rfb0810-681 rfb1010-152 lnk304 lnk305 lnk306 mdcm mdcm ccm mdcm ccm mdcm ccm 75 ns 75 ns 35 ns 75 ns 35 ns 75 ns 35 ns 11.86 k ? 11 v 15 70 120 160 175 225 280 360 680 160 1200 210 1800 210 820 310 1200 310 820 390 1500 390 sbc2-681-211 - - - - - sbc6-152-451 rfb0807-681 rfb0807-122 rfb0810-182 rfb0810-821 rfb1010-122 rfb1010-821 rfb1010-152 lnk304 lnk305 lnk306 mdcm mdcm ccm mdcm ccm mdcm ccm 75 ns 75 ns 35 ns 75 ns 35 ns 75 ns 35 ns 15.29 k ? 13 v 24 50 120 160 175 225 280 360 680 130 1500 190 2200 180 1200 280 1500 280 1200 350 2200 360 sbc2-681-211 sbc4-152-221 sbc4-222-211 - sbc6-152-451 - sbc6-222-351 rfb0807-681 rfb0810-152 rfb0810-222 rfb0810-122 rfb1010-152 rfb1010-122 - lnk304 lnk305 lnk306 mdcm mdcm ccm mdcm ccm mdcm ccm 75 ns 75 ns 35 ns 75 ns 35 ns 75 ns 35 ns 25.6 k ? 22 v other standard components r bias :2 k ? , 1%, 1/8 w c bp : 0.1 f, 50 v ceramic c fb : 10 f, 1.25 v o d fb : 1n4005gp r z : 470 ? to 2 k ? , 1/8 w, 5%
AN-37 5 a 1/04 mur160 uf4005 byv26c fe1a stta10 6 stta10 6u us1j 600 600 600 600 600 600 600 1 1 1 1 1 1 1 50 75 30 35 20 20 75 leaded leaded leaded leaded leaded smd smd vishay vishay vishay/philips vishay st microelectronics st microelectronics vishay part no. v rrm i f t rr (v) (a) (ns) package manufacturer t able 4. components quick select for buck-boost converters. v out i out(max) hi rms (ma) tokin coilcraft lnk30x mode diode t rr r fb v z 5 120 160 175 225 280 360 680 220 680 230 680 340 680 320 680 440 680 430 sbc2-681-211 sbc2-681-211 sbc3-681-361 sbc4-681-431 sbc4-681-431 sbc4-681-431 rfb0807-681 rfb0807-681 rfb0810-681 rfb0810-681 rfb0810-681 rfb0810-681 lnk304 lnk305 lnk306 mdcm ccm mdcm ccm mdcm ccm 75 ns 35 ns 75 ns 35 ns 75 ns 35 ns 3.84 k ? 3.9 v inductor 12 70 120 160 175 225 280 360 680 180 1200 220 1800 210 820 320 1200 310 820 410 1800 410 sbc2-681-211 - - - - - - rfb0807-681 rfb1010-122 rfb0807-182 rfb0807-821 rfb0810-122 rfb0810-821 rfb1010-182 lnk304 lnk305 lnk306 mdcm mdcm ccm mdcm ccm mdcm ccm 75 ns 75 ns 35 ns 75 ns 35 ns 75 ns 35 ns 11.86 k ? 11 v 15 50 120 160 175 225 280 360 680 180 1500 220 2200 220 1000 320 1500 320 1200 400 2200 410 sbc2-681-211 sbc3-152-251 sbc4-222-211 sbc4-102-291 sbc4-152-251 - sbc6-222-351 rfb0807-681 rfb0807-152 rfb0810-222 rfb0810-102 rfb0810-152 rfb0810-122 rfb1010-222 lnk304 lnk305 lnk306 mdcm mdcm ccm mdcm ccm mdcm ccm 75 ns 75 ns 35 ns 75 ns 35 ns 75 ns 35 ns 15.29 k ? 13 v 24 35 120 160 175 225 280 360 680 180 2200 210 3300 210 1800 300 2200 290 1800 370 3300 410 sbc2-681-211 sbc3-222-191 sbc4-332-161 - sbc4-222-211 - - rfb0807-681 rfb0810-222 rfb0810-332 rfb0810-182 rfb1010-222 rfb1010-182 - lnk304 lnk305 lnk306 mdcm mdcm ccm mdcm ccm mdcm ccm 75 ns 75 ns 35 ns 75 ns 35 ns 75 ns 35 ns 25.6 k ? 22 v t able 5. list of ultra-fast diodes suitable for use as the freewheeling diode. other standard components r bias :2 k ? , 1%, 1/8 w c bp : 0.1 f, 50 v ceramic c fb : 10 f, 1.25 v o d fb : 1n4005gp r z : 470 ? to 2 k ? , 1/8 w, 5%
AN-37 6 a 1/04 reference schematic and key t able 6. linkswitch-tn operation. linkswitch-tn circuit design linkswitch-tn operation the basic circuit configuration for a buck converter using linkswitch-tn is shown in figure 1a. v o v in pi-3784-121603 + + fb bp s d linkswitch-tn = mosfet enabled = mosfet disabled - cycle skipped normal operation auto restart at beginning of each cycle the feedback (fb) pin is sampled. if i fb < 49 a then next cycle occurs if i fb > 49 a then next switching cycle is skipped high load ?few cycles skipped low load ?many cycles skipped if no feedback (i fb < 49 a) for > 50 ms then output switching is disabled for approximately 800 ms. to regulate the output, an on/off control scheme is used as illustrated in table 6. as the decision to switch is made on a cycle by cycle basis, the resultant power supply has extremely good transient response and removes the need for control loop compensation components. if no feedback is received for 50 ms then the supply enters auto restart. i fb < 49 a, > 50 ms = auto restart auto restart = 50 ms on / 800 ms off 50 ms 800 ms pi-3768-121603 i d no no yes no is i fb >49 a? no yes yes no pi-3767-121903
AN-37 7 a 1/04 operating mode comparison of ccm and mdcm operating modes mdcm ccm operating description inductor freewheeling diode linkswitch-tn efficiency overall inductor current falls to zero during t off , borderline between mdcm and ccm when t idle = 0. current flows continuously in the inductor for the entire duration of a switching cycle. t able 7. comparison of mostly discontinuous conduction (mdcm) and continuous conduction (ccm) modes of operation. lower cost lower value, smaller size. higher cost higher value, larger size. lower cost 75 ns ultra-fast reverse recovery type ( 35 ns for ambient >70 c). higher cost 35 ns ultra-fast recovery type required. potentially higher cost may require larger device to deliver required output current?epends on required output current. potentially lowest cost may allow smaller device to deliver required output current?epends on required output current. higher efficiency lower switching losses. lower efficiency higher switching losses. t ypically lower cost t ypically higher cost to allow direct sensing of the output voltage, without the need for a reference (zener diode or reference ic), the fb pin voltage is tightly toleranced over the entire operating temperature range. for example, this allows a 12 v design with an overall output tolerance of 10%. for higher performance, an opto- coupler can be used with a reference as shown in table 2. since the optocoupler just provides level shifting, it does not need to be safety rated or approved. the use of an optocoupler also allows flexibility in the location of the device, for example it allows a buck converter configuration with the linkswitch-tn in the low side return rail, reducing emi as the source pins and connected components are no longer part of the switching node. selecting the topology if possible use the buck topology, the buck topology maximizes the available output power from a given linkswitch-tn and inductor value. also, the voltage stress on the power switch and freewheeling diode, and the average current through the output inductor are slightly lower in the buck topology as compared to the buck-boost topology. selecting the operating mode ?mdcm and ccm operation at the start of a design, select between mostly discontinuous conduction mode (mdcm) and continuous conduction mode (ccm) as this decides the selection of the linkswitch-tn device, freewheeling diode and inductor. for maximum output current select ccm, for all other cases mdcm is recommended. overall, select the operating mode and components to give the lowest overall solution cost. table 7 summarizes the trade-offs between the two operating modes. additional differences between ccm and mdcm include better transient response for dcm and lower output ripple (for same capacitor esr) for ccm. however these differences, at pi-3769-121803 i l t on t off t idle t i o pi-3770-121503 i l t on t off t i o
AN-37 8 a 1/04 the low output currents of linkswitch-tn applications, are normally not significant. the conduction mode ccm or mdcm of a buck or buck- boost converter primarily depends on input voltage, output voltage, output current and device current limit. the input voltage, output voltage and output current are fixed design parameters therefore the linkswitch-tn (current limit) is the only design parameter that sets the conduction mode. the phrase ?ostly discontinuous?is used as with on-off control, since a few switching cycles may exhibit continuous inductor current, the majority of the switching cycles will be in the discontinuous conduction mode. a design can be made fully discontinuous but that will limit the available output current, making the design less cost effective. step-by-step design procedure step 1. determine system requirements vac min , vac max , p o , v o , f l , determine the input voltage range from table 8. line frequency, f l : 50 or 60 hz, for half-wave rectification use f l /2. output voltage, v o : in volts. output power, p o : in watts. power supply efficiency, : 0.7 for a 12 v output, 0.55 for a 5 v output if no better reference data available. step 2. determine ac input stage the input stage comprises fusible resistor(s) input rectification diodes and line filter network. the fusible resistor should be chosen as flame proof and depending on the differential line input surge requirements, a wire wound type may be required. the fusible resistor(s) provides fuse safety, inrush current limiting and differential mode noise attenuation. for designs 1 w it is lower cost to use half-wave rectification, >1 w full wave rectification (smaller input capacitors). the emi performance of half wave rectified designs is improved by adding a second diode in the lower return rail. this provides emi gating (emi currents only flow when the diode is conducting) and also doubles differential surge withstand as the surge voltage is shared across two diodes. table 9 shows the recommended input stage based on output power for a universal input design while table 10 shows how to adjust the input capacitance for other input voltage ranges. p out 0.25 w 0.25-1 w > 1 w t able 9. recommended ac input stages for universal input. pi-3774-121603 + ac in r f1 l in d in1-4 c in2 c in1 ** pi-3773-121603 + ac in r f1 l in d in1 d in2 ** * c in2 c in1 pi-3772-121603 + ac in r f1 r f2 d in1 d in2 ** * c in2 c in1 pi-3771-121603 + ac in r f1 r f2 d in1 d in2 c in ** ** 85-265 vac input stage r f1 , r f2 : 100-470 ? , 0.5 w, fusible c in : 2.2 f, 400 v d in1 , d in2 : 1n4007, 1 a, 1000 v r f1 : 8.2 ? , 1 w fusible r f2 : 100 ? , 0.5 w, flame proof c in1 , c in2 : 3.3 f, 400 v each d in1 , d in2 : 1n4007, 1 a, 1000 v r f1 : 8.2 ? , 1 w fusible l in : 470 h-2.2 mh, 0.05 a-0.3 a c in1 , c in2 : 4 f/w out , 400 v each d in1 , d in2 : 1n4007, 1 a, 1000 v r f1 : 8.2 ? , 1 w fusible l in : 470 h-2.2 mh, 0.05 a-0.3 a c in1 , c in2 : 2 f/w out , 400 v each d in1 , d in2 : 1n4005, 1 a, 600 v *optional for improved emi and line surge performance. remove for designs requiring no impedance in return rail. **increase value to meet required differential line surge performance. comments input (vac) vac min vac max 100/115 85 132 230 195 265 universal 85 265 t able 8. standard worldwide input line voltage ranges. ac input half wave full wave v oltage (vac) rectification rectification 100/115 6-8 3-4 230 1-2 1 universal 6-8 3-4 t able 10. suggested total input capacitance values for different input voltage ranges. t otal capacitance c in(total) f/p out (c in1 + c in2 )
AN-37 9 a 1/04 step 5. select the output inductor tables 3 and 4 provide inductor values and rms current ratings for common output voltages and currents based on the calculations in the design spreadsheet. select the next nearest higher voltage and/or current above the required output specification. alternatively the pixls spreadsheet tool in the pi expert software design suite or appendix a can be used to calculate the exact inductor value (eq. a7) and rms current rating (eq. a20). it is recommended that the value of inductor chosen should be closer to l typ rather than 1.5 l typ due to lower dc resistance and higher rms rating. the lower limit of 680 h limits the maximum di/dt to prevent very high peak current values. tables 3 and 4 provide reference part numbers for standard inductors from two suppliers. (5) for linkswitch-tn designs the mode of operation is not dependent on the inductor value. the mode of operation is a function of load current and current limit of the chosen device, the inductor value merely sets the average switching frequency. figure 2 shows a typical standard inductor manufacturer? data sheet. the value of off-the-shelf ?rum core / dog bone / i core inductors will drop up to 20% in value as the current increases. the constant k l_tol in equation (a7) and the design spreadsheet adjusts for both this drop and the initial inductance value tolerance. for example if a 680 h, 360 ma inductor is required, referring to figure 2, the tolerance is 10% and an estimated 9.5% for the reduction in inductance at the operating current (approximately [0.36/0.38] 10). therefore the value of k l_tol = 1.195 (19.5%). if no data is available assume a k l_tol of 1.15 (15%). not all the energy stored in the inductor is delivered to the load, due to losses in the inductor itself. to compensate for this a loss step 3. determine minimum and maximum dc input voltages v min and v max based on ac input voltage calculate v max as (1) assuming that the value of input fusible resistor is small, the voltage drop across it can be ignored. assume bridge diode conduction time of t c = 3 ms if no other data available. derive minimum input voltage v min (2) if v min is 70 v then increase value of c in(total) . step 4. select linkswitch-tn device based on output current and current limit decide on operating mode - refer to table 7. for mdcm operation, the output current (i o ) should be less than or equal to half the value of the minimum current limit of the chosen device from the data sheet. (3) for ccm operation, the device should be chosen such that the output current i o , is more than 50%, but less than 80% of the minimum current limit i limit_min . (4) please see data sheet for linkswitch-tn current limit values. inductance and t olerance sbc3 series (sbc3- - ) 681-361 68010% 1.62 0.36 0.50 0.38 102-281 100010% 2.37 0.28 0.39 0.31 152-251 150010% 3.64 0.25 0.35 0.26 222-191 220010% 5.62 0.19 0.26 0.21 332-151 330010% 7.66 0.15 0.21 0.17 current rating f or 20 c rise current rating f or 40 c rise current rating f or value -10% inductance rdc rated current current (reference value) (w) (a) (a) l(mh/ at 10 khz max. ? t = 20 c ? t = 40 c l change rate -10% model pi-3783-121003 figure 2. example of standard inductor data sheet. vv p f t c min acmin o l c in total = ? () ? ? ? ? ? ? ? ? ? ? 2 2 1 2 2 () ii limit min o _ > ? 2 05 08 .. __ ? << ? iii limit min o limit min vv max acmax = ? 2 680 ll 1.5 l typ typ h <<< ?
AN-37 10 a 1/04 factor k loss is used. this has a recommended value of between 50% and 66% of the total supply losses as given by equation (5). for example, a design with an overall efficiency ( ) of 0.75 would have a k loss value of between 0.875 and 0.833. (6) step 6. select freewheeling diode for mdcm operation at t amb 70 c, select an ultra-fast diode with t rr 75 ns. at t amb >70 c, t rr 35 ns. for ccm operation, select an ultra-fast diode with t rr 35 ns. allowing 25% design margin for the freewheeling diode, (7) the diode must be able to conduct the full load current. thus (8) table 5 lists common freewheeling diode choices. step 7. select output capacitor the output capacitor should be chosen based on the output voltage ripple requirement. typically the output voltage ripple is dominated by the capacitor esr and can be estimated as: (9) where v ripple is the maximum output ripple specification and i limit is the linkswitch-tn current limit. the capacitor esr value should be specified approximately at the switching frequency of 66 khz. capacitor values above 100 f are not recommended as they can prevent the output voltage from reaching regulation during the 50 ms period prior to auto-restart. if more capacitance is required, then a soft-start capacitor should be added (see other information section). step 8. select the feedback resistors the values of r fb and r bias are selected such that at the regulated output voltage, the voltage on the feedback pin (v fb ) is 1.65 v. this voltage is specified for a feedback pin current (i fb ) of 49 a. let the value of r bias = 2 k ? ; this biases the feedback network at a current of 0.8 ma. hence the value of r fb is given by (10) step 9. select the feedback diode and capacitor for the feedback capacitor, use a 10 f general purpose electrolytic capacitor with a voltage rating 1.25 v o . for the feedback diode, use a glass passivated 1n4005gp or 1n4937gp device with a voltage rating of 1.25 v max . step 10. select bypass capacitor use 0.1 f, 50 v ceramic capacitor. step 11. select pre-load resistor for direct feedback designs if the minimum load <3 ma then calculate r pl = v o / 3 ma. other information startup into non-resistive loads if the total system capacitance is >100 f or the output voltage is >12 v, then the output may fail to reach regulation during start-up. this may also be true when the load is not resistive, for example the output is supplying a motor or fan. to increase the startup time, a soft-start capacitor can be added across the feedback resistor, as shown in figure 3. the value of this soft-start capacitor is typically in the range of 0.47 f to 47 f with a voltage rating of 1.25 v o . figure 4 shows the effect of c ss used on a 12 v, 150 ma design driving a motor load. v o v in c ss r fb pi-3775-121003 + + fb bp s d linkswitch-tn vv piv max > ? 125 . ii fo > ? 125 . esr v i max ripple limit = r vv v r i vv r vir v fb ofb fb bias fb ofbb ias fb fb bias o = ? + = ? () ? + ? () = ? () ? 165 2 1 748 . . v k v ? figure 3. example schematic showing placement of soft-start capacitor. kto loss = ? ? () ? ? ? ? ? ? ? ? () ? ? ? ? ? ? 1 1 2 1 21 3 ?
AN-37 11 a 1/04 figure 4. example of using soft-start capacitor to enable driving a 12 v, 0.15 a motor load. all measurements were made at 85 vac (worst case condition). +7 v rtn -5 v v in 5v1 6v8 pi-3776-121003 + fb bp s d linkswitch-tn figure 5. example circuit ?generating dual output voltages. optional see text r sense r fb 300 ? r bias 2 k ? i o c o l c sense d fw vr fb d fb v in pi-3795-122403 + fb bp s d linkswitch-tn figure 6. high-side buck-boost constant current output configuration. 0 2.5 5 2 8 6 10 4 time (s) 12 14 0 -2 voltage (v) no soft-start capacitor. output never reaches regulation (in auto-restart). pi-3785-010504 0 2.5 5 2 8 6 10 4 time (s) 12 14 0 -2 voltage (v) soft-start capacitor value too small ?output still fails to reach regulation before auto-restart. pi-3786-010504 correct value of soft-start capacitor ?output reaches regulation before auto-restart. 0 2.5 5 2 8 6 10 4 time (s) 12 14 0 -2 voltage (v) pi-3787-010503 generating negative and positive outputs in appliance applications there is often a requirement to generate both an ac line referenced positive and negative output. this can be accomplished using the circuit in figure 5. the two zener diodes have a voltage rating close to the required output voltage for each rail and ensure that regulation is maintained when one rail is lightly and the other heavily loaded. the linkswitch-tn circuit is designed as if it were a single output voltage with an output current equal to the sum of both outputs. the magnitude sum of the output voltages in this example being 12 v. constant current circuit configuration (led driver) the circuit shown in figure 5 is ideal for driving constant current loads such as leds. it uses the tight tolerance and temperature stable feedback pin of linkswitch-tn as the reference to provide an accurate output current. to generate a constant current output, the average output current is converted to a voltage by resistor r sense and capacitor c sense and fed into the feedback pin via r fb and r bias . with the values of r bias and r fb as shown, the value of r sense should be chosen to generate a voltage drop of 2 v at the
AN-37 12 a 1/04 figure 7. inductor voltage and inductor current of a buck converter in dcm. pi-3778-121803 i limit v in -v o v o v l i l t on t off t idle i o t t required output current. capacitor c sense filters the voltage across r sense , which is modulated by inductor ripple current. the value of c sense should be large enough to minimize the ripple voltage, especially in mdcm designs. a value of c sense is selected such that the time constant (t) of r sense and c sense is greater than 20 times that of the switching period (15 s). the peak voltage seen by c sense is equal to r sense i limit(max) . the output capacitor is optional; however with no output capacitor the load will see the full peak current (i limit ) of the selected linkswitch-tn . increase the value of c o (typically in the range of 100 nf to 10 uf) to reduce the peak current to an acceptable level for the load. if the load is disconnected, feedback is lost and the large output voltage which results may cause circuit failure. to prevent this, a second voltage control loop, d fb and vr fb , can be added as shown if figure 6. this also requires that c o is fitted. the voltage of the zener is selected as the next standard value above the maximum voltage across the led string when it is in constant current operation. the same design equations / design spreadsheet can be used as for a standard buck-boost design, with the following additional considerations. 1. v o = led v f number of leds per string 2. i o = led i f number of strings 3. lower efficiency estimate due to r sense losses (enter r sense into design spreadsheet as inductor resistance) 4. set r bias = 2 k ? and r fb = 300 ? 5. r sense = 2/i o 6. c sense = 20 (15 s/r sense ) 7. select c o based on acceptable output ripple current through the load 8. if the load can be disconnected or for additional fault protection, add voltage feedback components d fb and vr fb , in addition to c o . thermal environment to ensure good thermal performance, the source pin temperature should be maintained below 100 c, by providing adequate heatsinking. for applications with high ambient temperature (>50 c), it is recommended to build and test the power supply at the maximum operating ambient temperature, and ensure that there is adequate thermal margin. the figures for maximum output current provided in the data sheet correspond to an ambient temperature of 50 c, and may need to be thermally derated. also, it is recommended to use ultra fast ( 35 ns) low reverse recovery diodes at higher operating temperatures (>70 c). recommended layout considerations traces carrying high currents should be as short in length and thick in width, as possible. these are the traces which connect the input capacitor, linkswitch-tn , inductor, freewheeling diode and the output capacitor. most off-the-shelf inductors are drum core inductors or dog- bone inductors. these inductors do not have a good closed magnetic path, and are a source of significant magnetic coupling. they are a source of differential mode noise and for this reason, they should be placed as far away as possible from the ac input lines. appendix a calculations for inductor value for buck and buck- boost topologies there is a minimum value of inductance that is required to deliver the specified output power, regardless of line voltage and operating mode. as a general case, figure 7 shows the inductor current in discontinuous conduction mode (dcm). the following expressions are valid for both ccm as well as dcm operation. there are three unique intervals in dcm as can be seen from figure 7. interval t on is when the linkswitch-tn is on and the freewheeling diode is off. current ramps up in the inductor from an initial value of zero. the peak current is the current limit i limit of the device. interval t off is when the linkswitch-tn is off and the freewheeling diode is on. current ramps down to zero during this interval. interval t idle is when both the linkswitch-tn and freewheeling diode are off, and the inductor current is zero.
AN-37 13 a 1/04 ? it i v l t off ripple o min off () == ? ii i initial limit min ripple = ? _ ? it i vvv l t iiit for ccm ii t for dcm on ripple min ds o min on ripple limit min o idle ripple limit min idle () == ?? ? = ?? () = () => () 20 0 _ _ , i t ii il vvv ii il v o sw max limit min initial ripple min min ds o limit min initial ripple min o = ? + () ? ?? + ? + () ? ? ? ? ? ? ? ? ? ? ? ? ? 1 1 2 1 2 _ _ _ i t iit iitt o sw max limit min initial on limit min initial off idle = ? + () ? + ? + () ? + ? ? ? ? ? ? ? ? ? 1 1 2 1 2 0 _ _ _ l vi v v v iifsvv min oo min ds o limit min initial min min ds = ?? () ??? () ? () ??? () 2 22 _ l k vi k vvv iifsvv typ ltol oo loss min ds o limit min initial min min ds = ?? ? ? ? ? ? ? ? ??? () ? () ??? () 2 22 _ _ pli i fs vv vvv k k o max typ limit min initial min min ds min ds o loss l tol __ _ = ?? ? () ? ? ? ?? ? 1 2 22 in ccm this idle state does not exist and thus t idle = 0. neglecting the forward voltage drop of the freewheeling diode, we can express the current swing at the end of interval t on in a buck converter as (a1) where i ripple = inductor ripple current i limit_min = minimum current limit v min = minimum dc bus voltage v ds = on state drain to source voltage drop v o = output voltage l min = minimum inductance similarly, we can express the current swing at the end of interval t off as (a2) the initial current through the inductor at the beginning of each switching cycle can be expressed as (a3) the average current through the inductor over one switching cycle is equal to the output current i o . this current can be expressed as (a4) where i o = output current. t sw_max = the switching interval corresponding to minimum switching frequency fs min . substituting for t on and t off from equations (a1) and (a2) we have (a5) (a6) this however does not account for the losses within the inductor (resistance of winding and core losses) and the freewheeling diode, which will limit the maximum power delivering capability and thus reduce the maximum output current. the minimum inductance must compensate for these losses in order to deliver specified full load power. an estimate of these losses can be made by estimating the total losses in the power supply, and then allocating part of these losses to the inductor and diode. this is done by the loss factor k loss which increases the size of the inductor accordingly. furthermore, typical inductors for this type of application are bobbin core or dog bone chokes. the specified current rating refer to a temperature rise of 20 c or 40 c and to an inductance drop of 10%. we must incorporate an inductance tolerance factor k l_tol within the expression for minimum inductance, to account for this manufacturing tolerance. the typical inductance value thus can be expressed as (a7) where k loss is a loss factor, which accounts for the off-state total losses of the inductor. k l_tol is the inductor tolerance factor and can be between 1.1 and 1.2. a typical value is 1.15. with this typical inductance we can express maximum output power as (a8)
AN-37 14 a 1/04 similarly for buck-boost topology the expressions for l typ and p o_max are (a9) (a10) average switching frequency since linkswitch-tn uses an on-off type of control, the frequency of switching is non-uniform due to cycle skipping. we can average this switching frequency by substituting the maximum power as the output power in equation (a8). simplifying, we have (a11) similarly for buck-boost converter, simplifying equation (a9) we have (a12) calculation of rms currents the rms current value through the inductor is mainly required to ensure that the inductor is appropriately sized and will not over heat. also, rms currents through the linkswitch-tn and freewheeling diode are required to estimate losses in the power supply. assuming ccm operation, the initial current in the inductor in steady state is given by (a13) for dcm operation this initial current will be zero. ii i l rms sw rms d rms ___ =+ 22 i t itdt sw rms avg t sw on _ = () ? 1 0 2 i t it dt d rms avg t tt d on on off _ = () ? + 1 2 i t itit dt l rms avg t sw d avg _ = () + () () ? 1 0 2 it i it lswd () = () () t+ it i v l ttt dlim it min o on sw () = ? < _ , the current through the linkswitch-tn as a function of time is given by (a14) the current through the freewheeling diode as a function of time is given by (a15) (a16) and the current through the inductor as a function of time is given by (a17) from the definition of rms currents we can express the rms currents through the switch, freewheeling diode and inductor as follows (a18) (a19) (a20) since the switch and freewheeling diode currents fall to zero during the turn off and turn on intervals respectively, the rms inductor current is simplified to (a21) it i v l t d limit min o () = ?? < 00 , _ it t t don () =< 00 , pli i o max typ limit min initial __ () = ?? ? 1 2 22 ii v l t initial limit min o off = ?? _ iti vvv l ttt sw initial min ds o on () =+ ?? ? < ,0 fs vik li i k vvv vv avg oo ltol limit initial loss min ds o min ds = ??? ?? () ? ?? ? 2 22 _ fs vi li i k k k avg oo limit initial loss ltol loss = ?? ?? () ? 2 22 _ it t tt sw on on () =< 0 , l k vi k iifs typ l tol oo loss limit min initial min = ?? ? ? ? ? ? ? ? ? () ? 2 22 _ _
AN-37 15 a 1/04 p arameter buck buck-boost t able a1. circuit characteristics for buck and buck-boost topologies. l typ f avg i sw (t) linkswitch-tn current i d (t) diode forward current i l (t) inductor current max drain v oltage iti vvv l tt t it tt sw init min ds o on sw on () =+ ?? ? () => , , 0 it i v l tt t it i v l t it t t dlim it min o on dlim it min o don () = ?? > () = ?? < () = _ _ , , , 00 0 it i t it lswd () = () + () fs vi li i k k avg oo limit initial l l loss = ?? ?? () ? 2 22 _ iti vv l tt t it tt sw init min ds on sw on () =+ ? ? () => , , 0 it i t it lswd () = () + () v max vv max o + l k vi k vvv iifsvv typ l oo l loss min ds o limit min initial min min ds = ?? ? ? ? ? ? ? ? ??? () ? () ??? () 2 22 _ _ fs vik li i k vvv vv typ oo l limit initial l loss min ds o min ds = ??? ?? () ? ? ?? ? 2 2 _ l k vi k iifs typ l oo l loss limit min initial min = ?? ? ? ? ? ? ? ? ? () ? 2 22 _ _ it i v l tt t it i v l t it t t d limit min o on d limit min o don () = ?? > () = ?? < () = _ _ , , , 00 0 table a1 lists the design equations for important parameters using the buck and buck-boost topologies.
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