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L6712 L6712A
TWO-PHASE INTERLEAVED DC/DC CONTROLLER
1

Features
2 PHASE OPERATION WITH SYNCHRONOUS RECTIFIER CONTROL ULTRA FAST LOAD TRANSIENT RESPONSE INTEGRATED HIGH CURRENT GATE DRIVERS: UP TO 2A GATE CURRENT 3 BIT PROGRAMMABLE OUTPUT FROM 0.900V TO 3.300V OR WITH EXTERNAL REF. 0.9% OUTPUT VOLTAGE ACCURACY 3mA CAPABLE AVAILABLE REFERENCE INTEGRATED PROGRAMMABLE REMOTE SENSE AMPLIFIER PROGRAMMABLE DROOP EFFECT 10% ACTIVE CURRENT SHARING ACCURACY DIGITAL 2048 STEP SOFT-START CROWBAR LATCHED OVERVOLTAGE PROT. NON-LATCHED UNDERVOLTAGE PROT. OVERCURRENT PROTECTION REALIZED USING THE LOWER MOSFET'S RdsON OR A SENSE RESISTOR OSCILLATOR EXTERNALLY ADJUSTABLE AND INTERNALLY FIXED AT 150kHZ POWER GOOD OUTPUT AND INHIBIT FUNCTION PACKAGES: SO-28 & VFQFPN-36
Figure 1. Packages
SO28
VFQFPN-36 (6x6x1.0mm)
Table 1. Order Codes
Tube L6712D, SO L6712AD L6712Q, VFQFPN L6712AQ Package Tape & Reel L6712DTR, L6712ADTR L6712QTR, L6712AQTR

1.1 Applications HIGH CURRENT DC/DC CONVERTERS DISTRIBUTED POWER SUPPLY
2
Description
The device implements a dual-phase step-down controller with a 180 phase-shift between each phase
optimized for high current DC/DC applications. Output voltage can be programmed through the integrated DAC from 0.900V to 3.300V; programming the "111" code, an external reference from 0.800V to 3.300V is used for the regulation. Programmable Remote Sense Amplifier avoids use of external resistor divider and recovers losses along distribution line. The device assures a fast protection against load over current and Over / Under voltage.An internal crowbar is provided turning on the low side mosfet if Over-voltage is detected. Output current is limited working in Constant Current mode: when Under Voltage is detected, the device resets, restarting operation.
June 2005
Rev. 3 1/29
L6712A L6712
Figure 2. Block Diagram
OSC / INH SGND VCCDR BOOT1
LOGIC PWM ADAPTIVE ANTI CROSS CONDUCTION
BAND-GAP REFERENCE
2 PHASE OSCILLATOR
HS
UGATE1 PHASE1
PGOOD
PWM1
CURRENT CORRECTION
VID2 VID1 VID0
DAC
LOGIC AND PROTECTIONS
VCC VCCDR
CH1 OCP
LS
LGATE1 ISEN1
CH1 OCP
CH2 OCP
TOTAL CURRENT CURRENT AVG
CURRENT READING
PGNDS1 PGND
DIGITAL SOFT-START
CURRENT READING
PGNDS2 ISEN2
CURRENT CORRECTION
REF_IN/OUT
IDROOP
CH2 OCP
FBG
IFB_START
LOGIC PWM ADAPTIVE ANTI CROSS CONDUCTION
VPROG
LS
LGATE2
PHASE2 HS UGATE2 BOOT2
FBR REMOTE AMPLIFIER
PWM2
ERROR AMPLIFIER
Vcc
VSEN
DROOP FB
COMP
Vcc
Table 2. Absolute Maximum Ratings
Symbol VCC, VCCDR VBOOT-VPHASE VUGATE1-VPHASE1 VUGATE2-VPHASE2 LGATE1, PHASE1, LGATE2, PHASE2 to PGND VID0 to VID2 All other pins to PGND VPHASEx UGATEX Pins OTHER PINS Sustainable Peak Voltage. T<20ns @ 600kHz Maximum Withstanding Voltage Range Test Condition: CDF-AEC-Q100-002"Human Body Model" Acceptance Criteria: "Normal Performance" To PGND Boot Voltage Parameter Value 15 15 15 -0.3 to Vcc+0.3 -0.3 to 5 -0.3 to 7 26 1500 2000 Unit V V V V V V V V V
Table 3. Thermal Data
Symbol Rthj-amb Tmax Tstg Tj PMAX Parameter Thermal Resistance Junction to Ambient 4 layer PCB (2s2p) Maximum junction temperature Storage temperature range Junction Temperature Range Max power dissipation at Tamb = 25C SO28 60 150 -40 to 150 -40 to 125 2 VFQFPN36 30 150 -40 to 150 -40 to 125 3.5 Unit C/W C C C W
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Figure 3. Pin Connection (Top view)
PGOOD BOOT2 VID1 VID0 VID2 FBG 20 N.C. 19 18 17 16 15 14 13 12 11 10 1 N.C 2 BOOT1 3 N.C. 4 VCC 5 SGND 6 SGND 7 COMP 8 FB 9 DROOP N.C. FBR 21
LGATE1 VCCDR PHASE1 UGATE1 BOOT1 VCC SGND COMP FB DROOP REF_IN/OUT VSEN ISEN1 PGNDS1
1 2 3 4 5 6 7 8 9 10 11 12 13 14
28 27 26 25 24 23 22 21 20 19 18 17 16 15
PGND LGATE2 PHASE2 UGATE2 BOOT2 PGOOD VID2 VID1 VID0 FBR FBG OSC/INH/FAULT ISEN2 PGNDS2
UGATE2 PHASE2 LGATE2 PGND PGND LGATE1 VCCDR PHASE1 UGATE1
28 29 30 31 32 33 34 35 36
27
26
25
24
23
22
OSC N.C. ISEN2 PGNDS2 PGNDS1 ISEN1 VSEN REF_IN/OUT N.C.
VFQFPN-36 SO28 Corner Pin internally connected to the Exposed Pad.
Table 4. Electrical Characteristcs (VCC = 12V10%, TJ = 0C to 70C unless otherwise specified)
Symbol Vcc SUPPLY CURRENT ICC ICCDR IBOOTx POWER-ON Turn-On VCC threshold Turn-Off VCC threshold Turn-On VCCDR Threshold Turn-Off VCCDR Threshold OSCILLATOR AND INHIBIT fOSC INH dMAX Initial Accuracy Inhibit threshold Maximum duty cycle OSC = OPEN OSC = OPEN; Tj=0C to 125C ISINK=5mA L6712, OSC = OPEN: IDROOP=0 OSC = OPEN; IDROOP=70A L6712A, OSC = OPEN Vosc FAULT Ramp Amplitude Voltage at pin OSC OVP Active 4.75 135 127 0.5 72 30 85 80 40 90 3 5.0 5.25 150 165 178 kHz kHz V % % % V V VCC Rising; VCCDR=5V VCC Falling; VCCDR=5V VCCDR Rising VCC=12V VCCDR Falling VCC=12V 8.2 6.5 4.2 4.0 9.2 7.5 4.4 4.2 10.2 8.5 4.6 4.4 V V V V VCC supply current VCCDR supply current Boot supply current HGATEx and LGATEx open VCCDR=BOOTx=12V LGATEx open; VCCDR=12V HGATEx open; PHASEx to PGND; VCC=BOOTx=12V 7.5 1.5 0.5 10 3 1 12.5 4 1.5 mA mA mA Parameter Test Condition Min. Typ. Max. Unit
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Table 4. Electrical Characteristcs (continued) (VCC = 12V10%, TJ = 0C to 70C unless otherwise specified)
Symbol REFERENCE AND DAC VOUT (1) REF_IN/OUT Output Voltage Accuracy Reference Accuracy Current Capability Load Regulation VPROG / REF_IN/ Accuracy with external reference OUT REF_IN/OUT IVID VVID VIDIL VIDIH ERROR AMPLIFIER VOS_EA SR IFB_START VOS_RA SR IISEN1, IISEN2 IPGNDSx IISEN1, IISEN2 IDROOP GATE DRIVERS tRISE HGATE IHGATEx RHGATEx tRISE LGATE ILGATEx RLGATEx High Side Rise Time High Side Source Current High Side Sink Resistance Low Side Rise Time Low Side Source Current Low Side Sink Resistance BOOTx-PHASEx=10V; CHGATEx to PHASEx=3.3nF BOOTx-PHASEx=10V BOOTx-PHASEx=12V; VCCDR=10V; CLGATEx to PGNDx=5.6nF VCCDR=10V VCCDR=12V 0.7 1.5 15 2 2 30 1.8 1.1 1.5 2.5 55 30 ns A ns A Offset DC Gain Slew-Rate Start-up Current Offset DC Gain Slew Rate Bias Current Bias Current Bias Current at Over Current Threshold Droop Current ILOAD 0 ILOAD = 100% 47.5 VSEN = 10pF ILOAD = 0 45 45 80 DIFFERENTIAL CURRENT SENSING 50 50 85 0 50 55 55 90 1 52.5 A A A A A COMP=10pF FB=SGND; During Soft Start... VSEN = FBG 65 -8 80 15 8 FB = COMP -5 80 15 5 mV dB V/s A mV dB V/s Input impedance VID pull-up Current VID pull-up Voltage VID Input Levels VIDx =SGND VIDx = OPEN Input Low Input High 1.0 IREF = from 0 to 3mA VID="111"; REF_IN/OUT = 0.8V to 3.3V -2.0 400 5 3 0.4 VIDx See Table 5, VID "11x" VID = "110" VIDx See Table 5, VID "111" -0.9 -1.0 VOUT -5 3 5.0 2.0 VOUT 0.9 1.0 VOUT+5 % % mV mA mV % k A V V V Parameter Test Condition Min. Typ. Max. Unit
DIFFERENTIAL AMPLIFIER (REMOTE BUFFER)
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Table 4. Electrical Characteristcs (continued) (VCC = 12V10%, TJ = 0C to 70C unless otherwise specified)
Symbol PROTECTIONS PGOOD OVP UVP VPGOODL IPGOODH Upper Threshold Lower Threshold Over Voltage Threshold Under Voltage Trip PGOOD Voltage Low PGOOD Leakage VSEN Rising VSEN Falling VSEN Rising VSEN Falling IPGOOD = -4mA VPGOOD = 5V 108 84 115 55 112 88 122 60 115 92 130 65 0.4 1 % % % % V A Parameter Test Condition Min. Typ. Max. Unit
Note: 1. Output voltage is specified including Error Amplifier Offset in the trimming chain. Remote Amplifier is not included.
Table 5. Voltage Identification (VID) Codes.
VID2 1 1 1 1 0 0 0 0 VID1 1 1 0 0 1 1 0 0 VID0 1 0 1 0 1 0 1 0 Output Voltage (V) Ext. Ref. 0.900 1.250 1.500 1.715 1.800 2.500 3.300
Table 6. Pin Function
N. (*) SO 1 2 3 4 5 VFQFPN 33 34 35 36 2 LGATE1 VCCDR PHASE1 UGATE1 BOOT1 Name Description Channel 1 LS driver output. A little series resistor helps in reducing device-dissipated power. LS drivers supply: it can be varied from 5V to 12V buses. Filter locally with at least 1F ceramic cap vs. PGND. Channel 1 HS driver return path. It must be connected to the HS1 mosfet source and provides the return path for the HS driver of channel 1. Channel 1 HS driver output. A little series resistor helps in reducing device-dissipated power. Channel 1 HS driver supply. This pin supplies the relative high side driver. Connect through a capacitor (100nF typ.) to the PHASE1 pin and through a diode to VCC (cathode vs. boot). Device supply voltage. The operative supply voltage is 12V 10%. Filter with 1F (Typ.) capacitor vs. GND. All the internal references are referred to this pin. Connect it to the PCB signal ground. This pin is connected to the error amplifier output and is used to compensate the control feedback loop. This pin is connected to the error amplifier inverting input and is used to compensate the control feedback loop.
6 7 8 9
4 5,6 7 8
VCC SGND COMP FB
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Table 6. Pin Function (continued)
N. (*) SO 10 VFQFPN 9 DROOP Name Description A current proportional to the sum of the current sensed in both channel is sourced from this pin (50A at full load, 70A at the Constant Current threshold). Short to FB to implement the Droop effect: the resistor connected between FB and VSEN (or the regulated output) allows programming the droop effect. Otherwise, connect to GND directly or through a resistor (43k max) and filter with 1nF capacitor. In this last case, current information can be used for other purposes. Reference input/output. Filter vs. GND with 1nF ceramic capacitor (a total of 100nF capacitor is allowed). It reproduces the reference used for the regulation following VID code: when VID=111, the reference for the regulation must be connected on this pin. References ranging from 0.800V up to 3.300V can be accepted. Connected to the output voltage it is able to manage Over & Under-voltage conditions and the PGOOD signal. It is internally connected with the output of the Remote Sense Amplifier for Remote Sense of the regulated voltage. Connecting 1nF capacitor max vs. GND can help in reducing noise injection at this pin. If no Remote Sense is implemented, connect it directly to the regulated voltage in order to manage OVP, UVP and PGOOD. Channel 1 current sense pin. The output current may be sensed across a sense resistor or across the low-side mosfet RdsON. This pin has to be connected to the low-side mosfet drain or to the sense resistor through a resistor Rg. The net connecting the pin to the sense point must be routed as close as possible to the PGNDS net in order to couple in common mode any picked-up noise. Channel 1 Power Ground sense pin. The net connecting the pin to the sense point must be routed as close as possible to the ISEN1 net in order to couple in common mode any picked-up noise. Channel 2 Power Ground sense pin. The net connecting the pin to the sense point must be routed as close as possible to the ISEN2 net in order to couple in common mode any picked-up noise. Channel 2 current sense pin. The output current may be sensed across a sense resistor or across the low-side mosfet RdsON. This pin has to be connected to the low-side mosfet drain or to the sense resistor through a resistor Rg. The net connecting the pin to the sense point must be routed as close as possible to the PGNDS net in order to couple in common mode any picked-up noise. Oscillator pin. It allows programming the switching frequency of each channel: the equivalent switching frequency at the load side results in being doubled. Internally fixed at 1.24V, the frequency is varied proportionally to the current sunk (forced) from (into) the pin with an internal gain of 6kHz/A (See relevant section for details). If the pin is not connected, the switching frequency is 150kHz for each channel (300kHz on the load). The pin is forced high (5V Typ.) when an Over Voltage is detected; to recover from this condition, cycle VCC. Forcing the pin to a voltage lower than 0.6V, the device stops operation and enters the inhibit state. Remote sense amplifier inverting input. It has to be connected to the negative side of the load to perform programmable remote sensing through apposite resistors (see relative section). Remote sense amplifier non-inverting input. It has to be connected to the positive side of the load to perform programmable remote sensing through apposite resistors (see relative section).
11
11
REF_IN / OUT
12
12
VSEN
13
13
ISEN1
14
14
PGNDS1
15
15
PGNDS2
16
16
ISEN2
17
18
OSC/INH FAULT
18
20
FBG
19
21
FBR
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Table 6. Pin Function (continued)
N. (*) SO 20 to 22 VFQFPN 22 to 24 VID0-2 Name Description Voltage IDentification pins. These input are internally pulled-up. They are used to program the output voltage as specified in Table 1 and to set the PGOOD, OVP and UVP thresholds. Connect to GND to program a `0' while leave floating to program a `1'. This pin is an open collector output and is pulled low if the output voltage is not within the above specified thresholds and during soft-start. It cannot be pulled up above 5V. If not used may be left floating. Channel 2 HS driver supply. This pin supplies the relative high side driver. Connect through a capacitor (100nF typ.) to the PHASE2 pin and through a diode to VCC (cathode vs. boot). Channel 2 HS driver output. A little series resistor helps in reducing device-dissipated power. Channel 2 HS driver return path. It must be connected to the HS2 mosfet source and provides the return path for the HS driver of channel 2. Channel 2 LS driver output. A little series resistor helps in reducing device-dissipated power. LS drivers return path. This pin is common to both sections and it must be connected through the closest path to the LS mosfets source pins in order to reduce the noise injection into the device. Thermal pad connects the silicon substrate and makes a good thermal contact with the PCB to dissipate the power necessary to drive the external mosfets.Connect to the GND plane with several vias to improve thermal conductivity.
23
25
PGOOD
24
27
BOOT2
25 26 27 28
28 29 30 31, 32
UGATE2 PHASE2 LGATE2 PGND
PAD
THERMAL PAD
(*) Pin not reported in QFN column have to be considered as Not Connected, not internally bonded.
Figure 4. Reference Schematic
Vin GNDin
CIN VCCDR VCC
BOOT1 HS1 L1 UGATE1 PHASE1 LGATE1 ISEN1 Rg PGNDS1 Rg S2 S1 S0 VID2 VID1 VID0 REF_IN/OUT
BOOT2 UGATE2 PHASE2 LGATE2 ISEN2 Rg HS2 L2 COUT LOAD
LS1
LS2
L6712 L6712A
PGNDS2 PGND PGOOD COMP CF DROOP RF Rg
PGOOD
OSC / INH SGND
FB RFB VSEN FBR FBG
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3
Device Description
The device is an integrated circuit realized in BCD technology. It provides complete control logic and protections for a high performance dual-phase step-down converter optimized for high current DC/DC applications. It is designed to drive N-Channel Mosfets in a two-phase synchronous-rectified buck topology. A 180 deg phase shift is provided between the two phases allowing reduction in the input capacitor current ripple, reducing also the size and the losses. The output voltage of the converter can be precisely regulated, programming the VID pins, from 0.900 to 3.300V with a maximum tolerance of 0.9% over temperature and line voltage variations. The programmable Remote Sense Amplifier avoids the use of external resistor divider allowing recovering drops across distribution lines and also adjusting output voltage to different values from the available reference. The device provides an average current-mode control with fast transient response. It includes a 150kHz free-running oscillator externally adjustable through a resistor. The error amplifier features a 15V/s slew rate that permits high converter bandwidth for fast transient performances. Current information is read across the lower mosfets RdsON or across a sense resistor placed in series to the LS mos in fully differential mode. The current information corrects the PWM outputs in order to equalize the average current carried by each phase. Current sharing between the two phases is then limited at 10% over static and dynamic conditions unless considering the sensing element spread. Droop effect can be programmed in order to minimize output filter and load transient response: the function can be disabled and the current information available on the pin can be used for other purposes. The device protects against Over-Current, with an OC threshold for each phase, entering in constant current mode. Since the current is read across the low side mosfets, the device keeps constant the bottom of the inductors current triangular waveform. When an Under Voltage is detected the device resets with all mosfets OFF and suddenly re-starts. The device also performs a crowbar Over-Voltage protection that immediately latches the operations turning ON the lower driver and driving high the FAULT pin. 3.1 OSCILLATOR The switching frequency is internally fixed at 150kHz. Each phase works at the frequency fixed by the oscillator so that the resulting switching frequency at the load side results in being doubled. The internal oscillator generates the triangular waveform for the PWM charging and discharging with a constant current an internal capacitor. The current delivered to the oscillator is typically 25A (Fsw=150kHz) and may be varied using an external resistor (ROSC) connected between OSC pin and SGND or Vcc. Since the OSC pin is maintained at fixed voltage (Typ. 1.237V), the frequency is varied proportionally to the current sunk (forced) from (into) the pin considering the internal gain of 6KHz/A. In particular connecting it to SGND the frequency is increased (current is sunk from the pin), while connecting ROSC to Vcc=12V the frequency is reduced (current is forced into the pin), according to the following relationships: 1.237 kHz 7.422 10 ROSC vs. GND: F SW = 150 [ KHz ] + --------------- 6 ---------- = 150 [ kHz ] + ----------------------------- [ kHz ] R OSC R OSC [ K ] A kHz 6.457 10 12 - 1.237 ROSC vs. 12V: F SW = 150 [ KHz ] - ------------------------ 6 ---------- = 150 [ kHz ] - ----------------------------- [ kHz ] A R OSC [ K ] R OSC Forcing 25A into this pin, the device stops switching because no current is delivered to the oscillator
7 6
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Figure 5. ROSC vs. Switching Frequency
14000
800
Rosc(K ) vs. 12V
12000 10000 8000 6000 4000 2000 0 25 50 75 100 125 150
Rosc(K) vs. GND
700 600 500 400 300 200 100 0 150 250 350 450 550 650
Frequency (kHz)
Frequency (kHz)
3.2 DIGITAL TO ANALOG CONVERTER AND REFERENCE The built-in digital to analog converter allows the adjustment of the output voltage from 0.900V to 3.300V as shown in Figure 6. Different voltages can be reached simply changing the Remote Amplifier Gain that acts as a resistor divider (See relevant section). The internal reference is trimmed during production process to have an output voltage accuracy of 0.9% and a zero temperature coefficient around 70C including also error amplifier offset compensation. It is programmed through the voltage identification (VID) pins. These are inputs of an internal DAC that is realized by means of a series of resistors providing a partition of the internal voltage reference. The VID code drives a multiplexer that select a voltage on a precise point of the divider (see Figure 6). The DAC output is delivered to an amplifier obtaining the VPROG voltage reference (i.e. the set-point of the error amplifier). Internal pull-ups are provided (realized with a 5A current generator up to 3V typ.); in this way, to program a logic "1" it is enough to leave the pin floating, while to program a logic "0" it is enough to short the pin to SGND. The device offers a bi-directional pin REF_IN/OUT: the internal reference used for the regulation is usually available on this pin with 3mA of maximum current capability except when VID code 111 is programmed; in this case the device accepts an external reference through the REF_IN/OUT pin and regulates on it. When external reference is used, it must range from 0.800V up to 3.300V to assure proper functionality of the device. Figure 6 shows a block schematic of how the Reference for the regulation is managed when internal or external reference is used. The voltage identification (VID) pin configuration or the external reference provided also sets the powergood thresholds (PGOOD) and the Over/Under voltage protection (OVP/UVP) thresholds.
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Figure 6. Reference Management
BAND-GAP REFERENCE (1.235V) CONTROL LOGIC INTERNAL REFERENCE EXTERNAL REFERENCE
VIDx
ERROR AMPLIFIER
VID3 VID2 VID1 DAC DIGITAL SOFT-START
VPROG
REF_IN/OUT
FB
COMP
The output regulated voltage accuracy can be extracted from the following relationships (worst case condition):
V OS_RA ( 8mV ) V OUT_TOT_ACC [ % ] = V OUT_ACC [ % ] + K OS -------------------- 100 = ( 0.9% ) + K OS -------------------- 100 V OUT V OUT
(worst case with internal reference)
V PROG V OS_RA V OS_EA V OUT_TOT_ACC [ % ] = EXT_REF_Accuracy[%] + ------------------------------------ [%] + -------------------------- 100 + K OS -------------------- 100= EXT_REF REF_IN/OUT V OUT ( 8mV ) ( 5mV ) = EXT_REF_Accuracy[%] + ( 2.0% ) + -------------------------- 100 + K OS -------------------- 100 EXT_REF V out
(worst case with external reference) where VOS_RA and VOS_EA are the offsets related to the Error Amplifier and the Remote Amplifier respectively and KOS = 1+1/RA_Gain reflects the impact of the Remote Amplifier Gain (RA_Gain) on the regulation (see relevant section). A statistical analysis could consider applying the root-sum-square (RSS) method to calculate the precision since all the variables are statistically independent as follow:
V OUT_TOT_ACC [%] =
2 V OS_RA 2 ( V OUT_ACC [%] ) + K OS -------------------- 100 V OUT
(with internal reference)
2 2 2 V PROG V OS_EA V OS_RA 2 V OUT_TOT_ACC [%] = ( EXT_REF_Accuracy[%] ) + ------------------------------------ [%] + -------------------------- 100 + K OS -------------------- 100 REF_IN/OUT EXT_REF V OUT
(with external reference)
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3.3 DRIVER SECTION The integrated high-current drivers allow using different types of power MOS (also multiple MOS to reduce the RdsON), maintaining fast switching transition. The drivers for the high-side mosfets use BOOTx pins for supply and PHASEx pins for return. The drivers for the low-side mosfets use VCCDR pin for supply and PGND pin for return. A minimum voltage of 4.6V at VCCDR pin is required to start operations of the device. The controller embodies a sophisticated anti-shoot-through system to minimize low side body diode conduction time maintaining good efficiency saving the use of Schottky diodes in parallel to the LS mosfets. The dead time is reduced to few nanoseconds assuring that high-side and low-side mosfets are never switched on simultaneously: when the high-side mosfet turns off, the voltage on its source begins to fall; when the voltage reaches 2V, the low-side mosfet gate drive is applied with 30ns delay. When the lowside mosfet turns off, the voltage at LGATEx pin is sensed. When it drops below 1V, the high-side mosfet gate drive is applied with a delay of 30ns. If the current flowing in the inductor is negative, the source of high-side mosfet will never drop. To allow the turning on of the low-side mosfet even in this case, a watchdog controller is enabled: if the source of the high-side mosfet don't drop for more than 240ns, the low side mosfet is switched on so allowing the negative current of the inductor to recirculate. This mechanism allows the system to regulate even if the current is negative. The BOOTx and VCCDR pins are separated from IC's power supply (VCC pin) as well as signal ground (SGND pin) and power ground (PGND pin) in order to maximize the switching noise immunity. The separated supply for the different drivers gives high flexibility in mosfet choice, allowing the use of logic-level mosfet. Several combination of supply can be chosen to optimize performance and efficiency of the application. Power conversion is also flexible; 5V or 12V bus can be chosen freely. The peak current is shown for both the upper and the lower driver of the two phases in Figure 7. A 10nF capacitive load has been used. For the upper drivers, the source current is 1.9A while the sink current is 1.5A with VBOOT -VPHASE = 12V; similarly, for the lower drivers, the source current is 2.4A while the sink current is 2A with VCCDR = 12V. Figure 7. Drivers peak current: High Side (left) and Low Side (right)
CH3 = HGATE1; CH4 = HGATE2
CH3 = LGATE1; CH4 = LGATE2
3.4 CURRENT READING AND OVER CURRENT The current flowing trough each phase is read using the voltage drop across the low side mosfets RdsON or across a sense resistor (RSENSE) in series to the LS mosfet and internally converted into a current. The transconductance ratio is issued by the external resistor Rg placed outside the chip between ISENx and PGNDSx pins toward the reading points. The differential current reading rejects noise and allows to place sensing element in different locations without affecting the measurement's accuracy. The current reading
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circuitry reads the current during the time in which the low-side mosfet is on (OFF Time). During this time, the reaction keeps the pin ISENx and PGNDSx at the same voltage while during the time in which the reading circuitry is off, an internal clamp keeps these two pins at the same voltage sinking from the ISENx pin the necessary current (Needed if low-side mosfet RdsON sense is implemented to avoid absolute maximum rating overcome on ISENx pin). The proprietary current reading circuit allows a very precise and high bandwidth reading for both positive and negative current. This circuit reproduces the current flowing through the sensing element using a high speed Track & Hold transconductance amplifier. In particular, it reads the current during the second half of the OFF time reducing noise injection into the device due to the mosfet turn-on (See Figure 8-left). Track time must be at least 200ns to make proper reading of the delivered current. This circuit sources a constant 50A current from the PGNDSx pin: it must be connected through the Rg resistor to the ground side of the sensing element (See Figure 8-right). The two current reading circuitries use this pin as a reference keeping the ISENx pin to this voltage. The current that flows in the ISENx pin is then given by the following equation: R SENSE I PHASEx I ISENx = 50A + ------------------------------------------------ = 50A + I INFOx Rg Where RSENSE is an external sense resistor or the RdsON of the low side mosfet and Rg is the transconductance resistor used between ISENx and PGNDSx pins toward the reading points; IPHASEx is the current carried by the relative phase. The current information reproduced internally is represented by the second term of the previous equation as follow: R SENSE I PHASEx I INFOx = -----------------------------------------------Rg Since the current is read in differential mode, also negative current information is kept; this allow the device to check for dangerous returning current between the two phases assuring the complete equalization between the phase's currents. From the current information of each phase, information about the total current delivered (IFB = IINFO1 +IINFO2) and the average current for each phase (IAVG = (IINFO1 +IINFO2)/2 ) is taken. IINFOX is then compared to IAVG to give the correction to the PWM output in order to equalize the current carried by the two phases. Figure 8. Current reading timing (left) and circuit (right)
ILS1 LGATEx Rg ILS2 RSENSE ISENx IISENx IFB PGNDSx
Track & Hold
Rg
50A
The transconductance resistor Rg can be designed in order to have current information of 25A per phase at full nominal load; the over current intervention threshold is set at 140% of the nominal (IINFOx = 35A). According to the above relationship, the over current threshold (IOCPx) for each phase, which has to be placed at 1/2 of the total delivered maximum current, results:
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35A Rg I OCPx = -------------------------R SENSE I OCPx R SENSE Rg = -----------------------------------------35A
Since the device senses the output current across the low-side mosfets (or across a sense resistors in series with them) the device limits the bottom of the inductor current triangular waveform: an over current is detected when the current flowing into the sense element is greater than IOCPx (IINFOx > 35A). L6712 - Dynamic Maximum Duty Cycle Limitation The maximum duty cycle is limited as a function of the measured current and, since the oscillator frequency is fixed once programmed, imply a maximum on-time limitation as follow (where T is the switching period T=1/fSW and IOUT is the output current): R SENSE 0.80 - --------------------- I T = T = 0.80 T I FB = 0A - OUT 5.73k T ON,MAX = ( 0.80 - I FB 5.73k ) T = Rg T = 0.40 T I FB = 70A
This linear dependence has a value at zero load of 0.80*T and at maximum current of 0.40*T typical and results in two different behaviors of the device: Figure 9. TON Limited Operation
VOUT 0.80*VIN VOUT 0.80*VIN
T ON Limited Output characteristic
Resulting Output characteristic Desired Output characteristic and UVP threshold
0.40*VIN
0.40*VIN
IOCP=2*IOCPx (IDROOP=70A)
IOUT
IOCP=2*IOCPx (IDROOP=70A)
IOUT
a) Maximum output Voltage
b) TON Limited Output Voltage
TON Limited Output Voltage.
FOx
This happens when the maximum ON time is reached before the current in each phase reaches IOCPx (IIN< 35A).
Figure 9a shows the maximum output voltage that the device is able to regulate considering the TON limitation imposed by the previous relationship. If the desired output characteristic crosses the TON limited maximum output voltage, the output resulting voltage will start to drop after crossing. In this case, the device doesn't perform constant current limitation but only limits the maximum duty cycle following the previous relationship. The output voltage follows the resulting characteristic (dotted in Figure 9b) until UVP is detected or anyway until IFB = 70A. Constant Current Operation This happens when ON time limitation is reached after the current in each phase reaches IOCPx (IINFOx > 35A). The device enters in Quasi-Constant-Current operation: the low-side mosfets stays ON until the current read becomes lower than IOCPx (IINFOx < 35A) skipping clock cycles. The high side mosfets can be turned ON with a TON imposed by the control loop at the next available clock cycle and the device works in the
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L6712A L6712
usual way until another OCP event is detected. This means that the average current delivered can slightly increase also in Over Current condition since the current ripple increases. In fact, the ON time increases due to the OFF time rise because of the current has to reach the IOCPx bottom. The worst-case condition is when the ON time reaches its maximum value. When this happens, the device works in Constant Current and the output voltage decrease as the load increase. Crossing the UVP threshold causes the device to reset. Figure 10 shows this working condition. It can be observed that the peak current (Ipeak) is greater than the IOCPx but it can be determined as follow: V IN - Vout min V IN - Vout MIN I peak = I OCPx + ------------------------------------- Ton MAX = I OCPx + -------------------------------------- 0.40 T L L Where VoutMIN is the minimum output voltage (VID-40% as follow). The device works in Constant-Current, and the output voltage decreases as the load increase, until the output voltage reaches the under-voltage threshold (VoutMIN). The maximum average current during the Constant-Current behavior results: Ipeak - I OCPx I MAX,TOT = 2 I MAX = 2 I OCPx + ------------------------------------- 2 In this particular situation, the switching frequency results reduced. The ON time is the maximum allowed (TonMAX) while the OFF time depends on the application: Ipeak - I OCPx T OFF = L ------------------------------------V OUt Figure 10. Constant Current operation
Ipeak IMAX IOCPx
UVP Vout Droop effect
1 f = ----------------------------------------T ONmax + T OFF
TonMAX
TonMAX
(IDROOP=50A)
Iout IMAX,TOT
2*IOCPx (IDROOP=70A)
a) Maximum current for each phase
b) Output Characteristic
Over current is set anyway when IINFOx reaches 35A (IFB=70A). The full load value is only a convention to work with convenient values for IFB. Since the OCP intervention threshold is fixed, to modify the percentage with respect to the load value, it can be simply considered that, for example, to have on OCP threshold of 200%, this will correspond to IINFOx = 35A (IFB = 70A). The full load current will then correspond to IINFOx = 17.5A (IFB = 35A). Once the UVP threshold has been intercepted, the device resets with all power mosfets turned OFF. Another soft start is then performed allowing the device to recover from OCP once the over load cause has been removed. Crossing the UVP threshold causes the device to reset: all mosfets are turned off and a new soft start is
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L6712A L6712
then implemented allowing the device to recover if the over load cause has been removed.
L6712A - Fixed Maximum Duty Cycle Limitation The maximum duty cycle is fixed and constant with the delivered current. The device works in constant current operation once the OCP threshold has overcome. Refer to the above Constant Current section in which only the different value in the maximum duty has to be considered as follow:
V IN - Vout min V IN - Vout MIN I peak = I OCPx + ------------------------------------- Ton MAX = I OCPx + -------------------------------------- 0.85 T L L All the above reported relationships about the deliverable current once in quasi-constant current and constant current are still valid in this case. 3.5 REMOTE SENSE AMPLIFIER Remote Sense Amplifier is integrated in order to recover from losses across PCB traces and wiring in high current DC/DC converter remote sense of the regulated voltage is required to maintain precision in the regulation. The integrated amplifier is a low-offset error amplifier; external resistors are needed as shown in Figure 11 to implement a differential remote sense amplifier. Figure 11. Remote Sense Amplifier Connections
Reference
Reference ERROR AMPLIFIER
REMOTE AMPLIFIER
ERROR AMPLIFIER
IDROOP
REMOTE AMPLIFIER
IDROOP
FBR FBG
VSEN
DROOP
FB
COMP
FBR FBG
VSEN
DROOP
FB
COMP
R2 R1
Remote VOUT
R2 R1
Remote Ground
RFB
CF
RF
RFB
CF
RF
VOUT
RB used
RB Not Used
Equal resistors give to the resulting amplifier a unity gain: the programmed reference will be regulated across the remote load. To regulate output voltages different from the available references, the Remote Amplifier gain can be adjusted simply changing the value of the external resistors as follow (see Figure 11): V VSEN RA_Gain = ---------------------------------------------------------------------------------------- = R2 ------Remote_V OUT - Remote_GND R1 to regulate a voltage double of the reference, the above reported gain must be equal to 1/2. Modifying the Remote Amplifier Gain (in particular with values higher than 1) allows also to regulate voltages lower than the programmed reference. Since this Amplifier is connected as a differential amplifier, when calculating the offset introduced in the regulated output voltage, the "native" offset of the amplifier must be multiplied by the term KOS = [1+(1/RA_Gain)] because a voltage generator insisting on the non-inverting input represents the offset. If remote sense is not required, it is enough connecting RFB directly to the regulated voltage: VSEN becomes not connected and still senses the output voltage through the remote amplifier. In this case the use of the external resistors R1 and R2 becomes optional and the Remote Sense Amplifier can simply be connected as a "buffer" to keep VSEN at the regulated voltage (See Figure 11). Avoiding use of Remote Amplifier saves its offset in the accuracy calculation but doesn't allow remote sensing.
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3.6 INTEGRATED DROOP FUNCTION (Optional) Droop function realizes dependence between the regulated voltage and the delivered current (Load Regulation). In this way, a part of the drop due to the output capacitor ESR in the load transient is recovered. As shown in Figure 12, the ESR drop is present in any case, but using the droop function the total deviation of the output voltage is minimized. Connecting DROOP pin and FB pin together, forces a current IDROOP, proportional to the output current, into the feedback resistor RFB implementing the load regulation dependence. If RA_Gain is the Remote Amplifier gain, the Output Characteristic is then given by the following relationship (when droop enabled): R SENSE 1 1 V OUT = ------------------------ ( VID - R FB I DROOP ) = ------------------------ VID - R FB --------------------- I OUT RA_Gain RA_Gain Rg with a remote amplifier gain of 1/2, the regulated output voltage results in being doubled. The Droop current is equal to 50A at nominal full load and 70A at the OC intervention threshold, so the maximum output voltage deviation is equal to:
1 V FULL - POSITIVE - LOAD = - ------------------------ R 50A RA_Gain FB 1 V OC - INTERVENTION = - ------------------------ R 70A RA_Gain FB
Droop function is provided only for positive load; if negative load is applied, and then IINFOx<0, no current is sunk from the FB pin. The device regulates at the voltage programmed by the VID. If this effect is not desired, shorting DROOP pin to SGND, the device regulates as a Voltage Mode Buck converter. Figure 12. Load Transient response (Left) and DROOP pin connection (Right).
ESR DROP
REMOTE AMPLIFIER
Reference ERROR AMPLIFIER
VMAX VNOM VMIN DROOP PIN = GND DROOP PIN = FB PIN
R2 R1
Remote VOUT
IDROOP
FBR FBG
VSEN
DROOP
FB
COMP
R2 R1
Remote Ground
RFB
CF
RF
Short to GND if DROOP function is not implemented (Classic Voltage Mode).
3.7 MONITOR AND PROTECTIONS The device monitors through pin VSEN the regulated voltage in order to build the PGOOD signal and manage the OVP / UVP conditions.
PGOOD. Power good output is forced low if the voltage sensed by VSEN is not within 12% (Typ.) of the programmed value (RA_Gain=1). It is an open drain output and it is enabled only after the soft start is finished (2048 clock cycles after start-up). During Soft-Start this pin is forced low. UVP. If the output voltage monitored by VSEN drops below the 60% of the reference voltage for more than one clock period, the device turns off all mosfets and resets restarting operations with a new softstart phase (hiccup mode, see Figure 13). OVP. Enabled once VCC crosses the turn-ON threshold: when the voltage monitored by VSEN reaches 115% (min) of the programmed voltage (or the external reference) the controller permanently switches on both the low-side mosfets and switches off both the high-side mosfets in order to protect the load. The OSC/ FAULT pin is driven high (5V) and power supply (VCC) turn off and on is required to restart operations. Both Over Voltage and Under Voltage are active also during soft start (Under Voltage after than the
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reference voltage reaches 0.6V). The reference used in this case to determine the UV thresholds is the increasing voltage driven by the 2048 soft start digital counter while the reference used for the OV threshold is the final reference programmed by the VID pins or available on the REF_IN/OUT pin. Figure 13. UVP Protection & Hiccup Mode.
CH1=PGOOD; CH2=Vout; CH3=REF_OUT; CH4=Iout
3.8 SOFT START AND INHIBIT At start-up a ramp is generated increasing the loop reference from 0V to the final value programmed by VID in 2048 clock periods as shown in Figure 14. Once the soft start begins, the reference is increased: upper and lower Mosfets begin to switch and the output voltage starts to increase with closed loop regulation. At the end of the digital soft start, the Power Good comparator is enabled and the PGOOD signal is then driven high (See Figure 14). The Under Voltage comparator is enabled when the increasing reference voltage reaches 0.6V while OVP comparator is always active with a threshold equal to the +15%_min of the final reference. The Soft-Start will not take place, if both VCC and VCCDR pins are not above their own turn-on thresholds. During normal operation, if any under-voltage is detected on one of the two supplies the device shuts down. Forcing the OSC/INH pin to a voltage lower than 0.5V (Typ.) disables the device: all the power mosfets and protections are turned off until the condition is removed. Figure 14. Soft Start.
VCC=VCCDR
Turn ON threshold
VLGATEx
t
VOUT
t
PGOOD
t
2048 Clock Cycles
t
Timing Diagram
Acquisition: CH1=PGOOD; CH2=VOUT; CH3=REF_OUT
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3.9 INPUT CAPACITOR The input capacitor is designed considering mainly the input RMS current that depends on the duty cycle as reported in Figure 15. Considering the two-phase topology, the input RMS current is highly reduced comparing with a single-phase operation. It can be observed that the input RMS value is one half of the single-phase equivalent input current in the worst case condition that happens for D=0.25 and D=0.75. The power dissipated by the input capacitance is then equal to: P RMS = ESR ( I RMS )
2
Input capacitor is designed in order to sustain the ripple relative to the maximum load duty cycle. To reach the RMS value needed and also to minimize components cost, the input capacitance is realized by more than one physical capacitor. The equivalent RMS current is simply the sum of the single capacitor's RMS current. Input bulk capacitor must be equally divided between high-side drain mosfets and placed as close as possible to reduce switching noise above all during load transient. Ceramic capacitor can also introduce benefits in high frequency noise de coupling, noise generated by parasitic components along power path. Figure 15. Input RMS Current vs. Duty Cycle (D) and Driving Relationships.
Rms Current Normalized (IRMS/IOUT)
0.50
Single Phase
Dual Phase 0.25
I rms =
I OUT ----------- 2D ( 1 - 2D ) 2 I OUT ----------- ( 2D - 1 ) ( 2 - 2D ) 2
if if
D < 0.5 D > 0.5
Where D = VOUT/VIN
0.25 0.50 0.75 Duty Cycle (VOUT/VIN)
3.10 OUTPUT CAPACITOR The output capacitor is a basic component for the fast response of the power supply. Two-phase topology reduces the amount of output capacitance needed because of faster load transient response (switching frequency is doubled at the load connections). Current ripple cancellation due to the 180 phase shift between the two phases also reduces requirements on the output ESR to sustain a specified voltage ripple. Moreover, if DROOP function is enabled, bigger ESR can be used still keeping the same transient tolerances. In fact, when a load transient is applied to the converter's output, for first few microseconds the current to the load is supplied by the output capacitors. The controller recognizes immediately the load transient and increases the duty cycle, but the current slope is limited by the inductor value. The output voltage has a first drop due to the current variation inside the capacitor (neglecting the effect of the ESL): V OUT = I OUT ESR
A minimum capacitor value is required to sustain the current during the load transient without discharge
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it. The voltage drop due to the output capacitor discharge is given by the following equation: I OUT L V OUT = ------------------------------------------------------------------------------4 C OUT ( V In d max - V OUT ) Where DMAX is the maximum duty cycle value. The lower is the ESR, the lower is the output drop during load transient and the lower is the output voltage static ripple. 3.11 INDUCTOR DESIGN The inductance value is defined by a compromise between the transient response time, the efficiency, the cost and the size. The inductor has to be calculated to sustain the output and the input voltage variation to maintain the ripple current IL between 20% and 30% of the maximum output current. The inductance value can be calculated with this relationship: V IN - V OUT V OUT L = ----------------------------- -------------fs I L V IN Where FSW is the switching frequency, VIN is the input voltage and VOUT is the output voltage. Increasing the value of the inductance reduces the ripple current but, at the same time, reduces the converter response time to a load transient. The response time is the time required by the inductor to change its current from initial to final value. Since the inductor has not finished its charging time, the output current is supplied by the output capacitors. Minimizing the response time can minimize the output capacitance required. The response time to a load transient is different for the application or the removal of the load: if during the application of the load the inductor is charged by a voltage equal to the difference between the input and the output voltage, during the removal it is discharged only by the output voltage. The following expressions give approximate response time for I load transient in case of enough fast compensation network response: L I t application = ----------------------------V IN - V OUT L Itremoval = -------------V OUT
2
The worst condition depends on the input voltage available and the output voltage selected. Anyway the worst case is the response time after removal of the load with the minimum output voltage programmed and the maximum input voltage available. 3.12 MAIN CONTROL LOOP The system control loop topology depends on the DROOP pin connection: if connected to FB (droop function active) an Average Current Mode topology must be considered while, if connected to GND (droop function not active) a Voltage Mode topology must be considered instead. Anyway, the system control loop encloses the Current Sharing control loop to allow proper sharing to the inductor' currents. Each loop gives, with a proper gain, the correction to the PWMs in order to minimize the error in its regulation: the Current Sharing control loop equalize the currents in the inductors while the output voltage control loop fixes the output voltage equal to the reference programmed by VID (with or without the droop effect and with or without considering the Remote Amplifier Gain). Figure 16 reports the block diagram of the main control loop.
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Figure 16. Main Control Loop Diagram
+
L1 PWM1 IINFO2 IINFO1 L2 PWM2 ERROR AMPLIFIER 4/5 COMP ZF(S) REFERENCE PROGRAMMED BY VID FB RFB CO RO
1/5
1/5
CURRENT SHARING DUTY CYCLE CORRECTION
+
+ -
RA_Gain
D03IN1518
3.12.1Current Sharing (CS) Control Loop Active current sharing is implemented using the information from Trans conductance differential amplifier. A current reference equal to the average of the read current (IAVG) is internally built; the error between the read current and this reference is converted to a voltage with a proper gain and it is used to adjust the duty cycle whose dominant value is set by the error amplifier at COMP pin (See Figure 17). The current sharing control is a high bandwidth control loop allowing current sharing even during load transients. The current sharing error is affected by the choice of external components; choose precise Rg resistor (1% is necessary) to sense the current. The current sharing error is internally dominated by the voltage offset of Trans conductance differential amplifier; considering a voltage offset equal to 2mV across the sense resistor, the current reading error is given by the following equation: I READ 2mV ------------------- = --------------------------------------R SENSE I MAX I MAX Figure 17. Current Sharing Control Loop.
+
L1 PWM1
1/5
1/5
CURRENT SHARING DUTY CYCLE CORRECTION
IINFO2 IINFO1
+
COMP
PWM2
L2
D02IN1393
VOUT
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Where IREAD is the difference between one phase current and the ideal current (IMAX/2). For RSENSE = 4m and IMAX = 40A the current sharing error is equal to 2.5%, neglecting errors due to Rg and RSENSE mismatches. 3.12.2Average Current Mode (ACM) Control Loop (DROOP=FB) The average current mode control loop is reported in Figure 18. The current information IDROOP sourced by the DROOP pin flows into RFB implementing the dependence of the output voltage from the read current. The ACM control loop gain results (obtained opening the loop after the COMP pin): PWM Z F ( s ) ( R DROOP + RA_Gain Z P ( s ) ) G LOOP ( s ) = - ----------------------------------------------------------------------------------------------------------------------ZF ( s ) 1 ( Z P ( s ) + Z L ( s ) ) -------------- + 1 + ----------- R FB A(s) A ( s ) Where:

Rsense R DROOP = --------------------- R FB is the equivalent output resistance determined by the droop function; Rg ZP(s) is the impedance resulting by the parallel of the output capacitor (and its ESR) and the applied load Ro; ZF(s) is the compensation network impedance; ZL(s) is the parallel of the two inductor impedance; A(s) is the error amplifier gain; V IN 4 PWM = -- ------------------ is the ACM PWM transfer function where VOSC is the oscillator ramp amplitude 5 V OSC and has a typical value of 3V RA_Gain is the Remote Amplifier Gain.
Removing the dependence from the Error Amplifier gain, so assuming this gain high enough, the control loop gain results: V IN ZF ( s ) RA_Gain Z P ( s ) 4 - Rs G LOOP ( s ) = - -- ------------------ ----------------------------------- ------- + -------------------------------------------- 5 V OSC Z P ( s ) + Z L ( s ) Rg R FB Considering now that in the application of interest it can be assumed that Ro>>RL; ESR<21/29
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Figure 18. ACM Control Loop Gain Block Diagram (left) and Bode Diagram (right).
dB
ZF CF IDROOP RF DROOP
GLOOP
RFB VCOMP COMP FB VID L/2 d*VIN Cout ESR Rout VOUT RA_Gain
K
ZF(s)
PWM
LC
4 VIN 1 K= 5 VOSC RFB dB
Z
T
The ACM control loop gain is designed to obtain a high DC gain to minimize static error and cross the 0dB axes with a constant -20dB/dec slope with the desired crossover frequency T. Neglecting the effect of ZF(s), the transfer function has one zero and two poles. Both the poles are fixed once the output filter is designed and the zero is fixed by ESR and the Droop resistance. To obtain the desired shape an RF-CF series network is considered for the ZF(s) implementation. A zero at F=1/RFCF is then introduced together with an integrator. This integrator minimizes the static error while placing the zero in correspondence with the L-C resonance a simple -20dB/dec shape of the gain is assured (See Figure 18). In fact, considering the usual value for the output filter, the LC resonance results to be at frequency lower than the above reported zero.Compensation network can be simply designed placing Z= LC and imposing the cross-over frequency T as desired obtaining: R FB V OSC 5 L -R F = ---------------------------------- -- T --------------------------------------------------------V IN 4 R DROOP ------------------------ + ESR 2 RA_Gain L Co -2 C F = ------------------RF
3.12.3Voltage Mode (VM) Control Loop (DROOP = SGND) Disconnecting the DROOP pin from the Control Loop, the system topology becomes a Voltage Mode. The simplest way to compensate this loop still keeping the same compensation network consists in placing the RF-CF zero in correspondence with the L-C filter resonance. The loop gain becomes now: V IN ZP ( s ) Z (s) - FG LOOP ( s ) = - ------------------ -------------- ----------------------------------- RA_Gain V OSC R FB Z P ( s ) + Z L ( s ) 3.13 LAYOUT GUIDELINES Since the device manages control functions and high-current drivers, layout is one of the most important things to consider when designing such high current applications. A good layout solution can generate a benefit in lowering power dissipation on the power paths, reducing radiation and a proper connection between signal and power ground can optimize the performance of the control loops.
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Integrated power drivers reduce components count and interconnections between control functions and drivers, reducing the board space. Here below are listed the main points to focus on when starting a new layout and rules are suggested for a correct implementation.
Power Connections. These are the connections where switching and continuous current flows from the input supply towards the load. The first priority when placing components has to be reserved to this power section, minimizing the length of each connection as much as possible.
To minimize noise and voltage spikes (EMI and losses) these interconnections must be a part of a power plane and anyway realized by wide and thick copper traces. Figure 19. Power connections and related connections layout guidelines (same for both phases).
VIN
Rgate HGATEx PHASEx L LS D COUT CIN LOAD
SGND +VCC VCC LS D CIN PHASEx L COUT LOAD BOOTx CBOOTx HS
VIN
HS
Rgate LGATEx PGNDx
CVCC
a. PCB power and ground planes areas
b. PCB small signal components placement
The critical components, i.e. the power transistors, must be located as close as possible, together and to the controller. Considering that the "electrical" components reported in figure are composed by more than one "physical" component, a ground plane or "star" grounding connection is suggested to minimize effects due to multiple connections. Figure 19a shows the details of the power connections involved and the current loops. The input capacitance (CIN), or at least a portion of the total capacitance needed, has to be placed close to the power section in order to eliminate the stray inductance generated by the copper traces. Low ESR and ESL capacitors are required.
Power Connections Related. Figure 19b shows some small signal components placement, and how and where to mix signal and power ground planes. The distance from drivers and mosfet gates should be reduced as much as possible. Propagation delay times as well as for the voltage spikes generated by the distributed inductance along the copper traces are so minimized.
In fact, the further the mosfet is from the device, the longer is the interconnecting gate trace and as a consequence, the higher are the voltage spikes corresponding to the gate PWM rising and falling signals. Even if these spikes are clamped by inherent internal diodes, propagation delays, noise and potential causes of instabilities are introduced jeopardizing good system behavior. One important consequence is that the switching losses for the high side mosfet are significantly increased. For this reason, it is suggested to have the device oriented with the driver side towards the mosfets and the GATEx and PHASEx traces walking together toward the high side mosfet in order to minimize distance (see Figure 20). In addition, since the PHASEx pin is the return path for the high side driver, this pin must be connected directly to the High Side mosfet Source pin to have a proper driving for this mosfet. For the LS mosfets, the return path is the PGND pin: it can be connected directly to the power ground plane (if
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implemented) or in the same way to the LS mosfets Source pin. GATEx and PHASEx connections (and also PGND when no power ground plane is implemented) must also be designed to handle current peaks in excess of 2A (30 mils wide is suggested). Gate resistors of few ohms help in reducing the power dissipated by the IC without compromising the system efficiency. The placement of other components is also important: - The bootstrap capacitor must be placed as close as possible to the BOOTx and PHASEx pins to minimize the loop that is created. - Decoupling capacitor from VCC and SGND placed as close as possible to the involved pins. - Decoupling capacitor from VCCDR and PGND placed as close as possible to those pins. This capacitor sustains the peak currents requested by the low-side mosfet drivers. - Refer to SGND all the sensible components such as frequency set-up resistor (when present) and Remote Amplifier Divider. - Connect SGND to PGND plane on a single point to improve noise immunity. Connect at the load side (output capacitor) if Remote Sense is not implemented to avoid undesirable load regulation effect. - An additional 100nF ceramic capacitor is suggested to place near HS mosfet drain. This helps in reducing noise. - PHASE pin spikes. Since the HS mosfet switches in hard mode, heavy voltage spikes can be observed on the PHASE pins. If these voltage spikes overcome the max breakdown voltage of the pin, the device can absorb energy and it can cause damages. The voltage spikes must be limited by proper layout, the use of gate resistors, Schottky diodes in parallel to the low side mosfets and/or snubber network on the low side mosfets, to a value lower than 26V, for 20ns, at FSW of 600kHz max. - Boot Capacitor Extra Charge. Systems that do not use Schottky diodes in parallel to the LS mosfet might show big negative spikes on the phase pin. This spike can be limited as well as the positive spike but has an additional consequence: it causes the bootstrap capacitor to be over-charged. This extra-charge can cause, in the worst case condition of maximum input voltage and during particular transients, that boot-to-phase voltage overcomes the abs. max. ratings also causing device failures. It is then suggested in this cases to limit this extra-charge by: - adding a small resistor in series to the boot diode (one resistor can be enough for all the diodes if placed upstream the diode anode) - using low capacitance diodes. Figure 20. Device orientation (left) and sense nets routing (right).
To LS mosfet
(or sense resistor)
Towards HS mosfet
(30 mils wide)
Towards LS mosfet
(30 mils wide)
ST L6917
Towards HS mosfet
(30 mils wide)
To LS mosfet
(or sense resistor)
To regulated output
Sense Connections. Remote Amplifier: Place the external resistors near the device to minimize noise injection and refer to SGND. The connections for these resistors (from the remote load) must be routed as parallel nets in order to compensate losses along the output power traces and also to avoid the pick-up of any noise. Connecting these pins in points far from the load will cause a non-optimum load regulation, increasing output tol-
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erance. Current Reading: The Rg resistor has to be placed as close as possible to the ISENx and PGNDSx pins in order to limit the noise injection into the device. The PCB traces connecting these resistors to the reading point must be routed as parallel traces in order to avoid the pick-up of any noise. It's also important to avoid any offset in the measurement and to get a better precision, to connect the traces as close as possible to the sensing elements, dedicated current sense resistor or low side mosfet RdsON. Moreover, when using the low side mosfet RdsON as current sense element, the ISENx pin is practically connected to the PHASEx pin. DO NOT CONNECT THE PINS TOGETHER AND THEN TO THE HS SOURCE! The device won't work properly because of the noise generated by the return of the high side driver. In this case route two separate nets: connect the PHASEx pin to the HS Source (route together with HGATEx) with a wide net (30 mils) and the ISENx pin to the LS Drain (route together with PGNDSx). Moreover, the PGNDSx pin is always connected, through the Rg resistor, to the PGND: DO NOT CONNECT DIRECTLY TO THE PGND! In this case the device won't work properly. Route anyway to the LS mosfet source (together with ISENx net). Right and wrong connections are reported in Figure 21. Symmetrical layout is also suggested to avoid any unbalance between the two phases of the converter. Figure 21. PCB layout connections for sense nets. NOT CORRECT
VIA to GND plane To PHASE connection
CORRECT
To LS Drain and Source To HS Gate and Source
Wrong (left) and correct (right) connections for the current reading sensing nets.
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4
Package informations
Figure 22. SO-28 Mechanical Data & Package Dimensions
DIM. MIN. A a1 b b1 C c1 D E e e3 F L S 7.4 0.4 17.7 10 0.1 0.35 0.23
mm TYP. MAX. 2.65 0.3 0.49 0.32 0.5 45 (typ.) 18.1 10.65 1.27 16.51 7.6 1.27 0.291 0.016 0.697 0.394 0.004 0.014 0.009 MIN.
inch TYP. MAX. 0.104 0.012 0.019 0.013 0.020
OUTLINE AND MECHANICAL DATA
0.713 0.419 0.050 0.65 0.299 0.050
SO-28
8 (max.)
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Figure 23. VFQFPN-36 Mechanical Data & Package Dimensions
mm DIM. MIN. A A1 A2 A3 b D D2 E E2 e L ddd 0.180 5.875 1.750 5.875 1.750 0.450 0.350 0.800 TYP. 0.900 0.020 0.650 0.250 0.230 6.000 3.700 6.000 3.700 0.500 0.550 0.300 6.125 4.250 6.125 4.250 0.550 0.750 0.080 0.007 0.231 0.069 0.231 0.069 0.018 0.014 MAX. 1.000 0.050 1.000 MIN. 0.031 TYP. 0.035 MAX. 0.039 inch
OUTLINE AND MECHANICAL DATA
0.0008 0.0019 0.025 0.01 0.009 0.236 0.146 0.236 0.146 0.020 0.022 0.012 0.241 0.167 0.241 0.167 0.022 0.029 0.003 0.039
VFQFPN-36 (6x6x1.0mm) Very Fine Quad Flat Package No lead
7185332 F
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5
Revision History
Table 7. Revision History
Date
March 2004 June 2005
Revision
2 3 First Issue in EDOCS.
Description of Changes
Changed look and feel. Inserted "Boot Capacitor Extra Charge" paragraph to page 27.
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L6712A L6712
Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics. The ST logo is a registered trademark of STMicroelectronics. All other names are the property of their respective owners (c) 2005 STMicroelectronics - All rights reserved STMicroelectronics group of companies Australia - Belgium - Brazil - Canada - China - Czech Republic - Finland - France - Germany - Hong Kong - India - Israel - Italy - Japan Malaysia - Malta - Morocco - Singapore - Spain - Sweden - Switzerland - United Kingdom - United States of America www.st.com
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