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TM
MP1567
1.2A Synchronous Rectified Step-Down Converter
The Future of Analog IC Technology
TM
DESCRIPTION
The MP1567 is a 1.2A, 800KHz DC to DC converter designed for low voltage applications requiring high efficiency. Capable of providing output voltages as low as 0.9V from a 3.3V supply voltage, the MP1567 eliminates the need for a 5V rail, providing over 90% efficiency via synchronous rectification and eliminating heat issues in confined spaces. Soft-start operation protects internal circuitry from hard turn on issues. Switching at 800KHz reduces the size of external components and thereby reduces board space. The MP1567 includes cycle-by-cycle current limiting and under voltage lockout. Internal power switches combined with the tiny 10-pin MSOP or QFN packages provide a solution requiring a minimum of space.
FEATURES
* * * * * * * * * * * * * * * * * 1.2A Output Current Synchronous Rectified Internal 180m and 220m Power Switches VIN Range of 2.6V to 6V Over 90% Efficiency Zero Current Shutdown Mode Under Voltage Lockout Protection Soft-Start Operation Thermal Shutdown Internal Current Limit (Source & Sink) Tiny 10-Pin MSOP or QFN Packages Evaluation Boards Available SOHO Routers, PCMCIA Cards, Mini PCI Handheld Computers, PDAs Cell Phones Digital Video Cameras Small LCD Displays
APPLICATIONS
EVALUATION BOARD REFERENCE
Board Number EV0033 (MP1567DK) EV0059 (MP1567DK) EV0060 (MP1567DQ) Dimensions 2.5"X x 2.0"Y x 0.7"Z 2.5"X x 2.0"Y x 0.4"Z 2.5"X x 2.0"Y x 0.4"Z
"MPS" and "The Future of Analog IC Technology" are Trademarks of Monolithic Power Systems, Inc.
TYPICAL APPLICATION
INPUT 2.6V to 6V
2 1 IN EN BS SW 3 7 9
10nF
Efficiency vs Load Current
100 90
EFFICIENCY (%)
VOUT 1.8V/1.2A
VIN=3.3V
OFF ON OPEN IF NOT USED
10 6
80 70 60 50 40 30 20 10 0 10 VOUT=1.8V 100 LOAD CURRENT (mA) 1000 VIN=5V VIN=4V
MP1567
SS 5 4
FB BP 8
SGND PGND COMP
10nF
10nF
1nF
MP1567_TAC_S01
MP1567_EC01
MP1567 Rev. 2.3 1/3/2006
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MP1567 - 1.2A SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER
PACKAGE REFERENCE
TOP VIEW
TOP VIEW
BS 1 2 3 4 5 10 9 8 7 6 EN BP COMP FB SS
BS IN SW PGND SGND
1 2 3 4 5
10 9 8 7 6
EN BP COMP FB SS
IN SW PGND SGND
MP1567_PD01_MSOP10
EXPOSED PAD ON BACKSIDE
MP1567_PD02_QFN10
Part Number* MP1567DK *
Package MSOP10
Temperature -40C to +85C
Part Number** MP1567DQ
Package QFN10 (3mm x 3mm)
Temperature -40C to +85C
For Tape & Reel, add suffix -Z (eg. MP1567DK-Z) For Lead Free, add suffix -LF (eg. MP1567DK-LF-Z)
** For Tape & Reel, add suffix -Z (eg. MP1567DQ-Z)
For Lead Free, add suffix -LF (eg. MP1567DQ-LF-Z)
ABSOLUTE MAXIMUM RATINGS (1)
Input Supply Voltage VIN ............................. 6.5V SW Voltage VSW ................... -0.3V to VIN + 0.3V BS to SW Voltage .........................-0.3V to +6V Voltage at All Other Pins ...............-0.3V to +6V Storage Temperature.............. -55C to +150C
Thermal Resistance
(3)
MSOP10 ................................ 150 ..... 65... C/W 3x3 QFN10 ............................. 50 ...... 12... C/W
Notes: 1) Exceeding these ratings may damage the device. 2) The device is not guaranteed to function outside of its operating conditions. 3) Measured on approximately 1" square of 1 oz copper.
JA
JC
Recommended Operating Conditions
(2)
Input Supply Voltage VIN ....................2.6V to 6V Output Voltage VOUT ........................0.9V to 4.5V Operating Temperature............. -40C to +85C
ELECTRICAL CHARACTERISTICS
VIN = 5V, TA = +25C, unless otherwise noted.
Parameter Input Voltage Range Input Undervoltage Lockout Input Undervoltage Lockout Hysteresis Shutdown Supply Current Operating Supply Current BP Voltage EN Input Low Voltage EN Input High Voltage EN Hysteresis EN Input Bias Current Oscillator Switching Frequency Maximum Duty Cycle Minimum On Time Symbol VIN Condition Min 2.6 Typ 2.2 100 0.5 1.2 2.4 1.5 100 1 fSW DMAX tON 800 VFB = 0.7V 85 200 Max 6 Units V V mV A mA V V V mV A KHz % ns
VBP VIL VHL
VEN 0.3V VEN > 2V, VFB = 1.1V VIN = 2.6 to 6V
1.0 1.8 0.4
MP1567 Rev. 2.3 1/3/2006
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TM
MP1567 - 1.2A SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS (continued)
VIN = 5V, TA = +25C, unless otherwise noted.
Parameter Error Amplifier Voltage Gain Transconductance COMP Maximum Output Current FB Regulation Voltage FB Input Bias Current Soft-Start Soft-Start Current Output Switch On-Resistance Switch On Resistance Synchronous Rectifier On Resistance Switch Current Limit (Source) Synchronous Rectifier Current Limit (Sink) Thermal Shutdown Symbol AVEA GEA VFB IFB ISS VIN = 5V VIN = 3V VIN = 5V VIN = 3V 1.5 875 FB = 0.9V Condition Min Typ 400 300 30 905 -100 2 265 330 220 270 2.0 350 160 Max Units V/V A/V A mV nA A m m m m A mA C
935
TYPICAL PERFORMANCE CHARACTERISTICS
VIN = 3.3V, VOUT = 1.8V, TA = +25C, unless otherwise noted. Current Limit vs. Load Transient 0.1A to 1A Load Step Duty Cycle
2.5 2.0 1.5 1.0 0.5 0
ILOAD 0.5A/div. VOUT 50mV/div.
CURRENT LIMIT (A)
VSS 1V/div. VOUT 1V/div. IIN 0.5A/div.
0
20 40 60 DUTY CYCLE (%)
80
MP1567-TPC01 MP1567-TPC02
2ms/div.
MP1567-TPC03
Output Short Circuit
IOUT = 1.2A
Steady State
VOUT 1V/div.
VSW 2V/div.
VSS 1mV/div. IINDUCTOR 1A/div. IINDUCTOR 1A/div.
2ms/div.
MP1567-TPC04 MP1567-TPC05
MP1567 Rev. 2.3 1/3/2006
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MP1567 - 1.2A SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER
PIN FUNCTIONS
Pin# 1 2 3 4 5 6 7 8 9 10 Name BS IN SW PGND SGND SS FB COMP BP EN Function Power Switch Boost. BS powers the gate of the high-side N-Channel power MOSFET switch. Connect a 10nF or greater capacitor between BS and SW. Internal Power Input. IN supplies the power to the MP1567 through the internal LDO regulator. Bypass IN to PGND with a 10F or greater capacitor. Connect IN to the input source voltage. Output Switching Node. SW is the source of the high-side N-Channel switch and the drain of the low-side N-Channel switch. Connect the output LC filter between SW and the output. Power Ground. PGND is the source of the N-Channel MOSFET synchronous rectifier. Connect PGND to SGND as close to the MP1567 as possible. Signal Ground. Soft-Start Input. Place a capacitor from SS to SGND to set the soft-start period. The MP1567 sources 2A from SS to the soft-start capacitor at start up. As the voltage at SS rises, the feedback threshold voltage increases to limit inrush current at start up. Feedback Input. FB is the inverting input of the internal error amplifier. Connect a resistive voltage divider from the output voltage to FB to set the output voltage. Compensation Node. COMP is the output of the error amplifier. Connect a series RC network to compensate the regulation control loop. Internal 2.4V Regulator Bypass. Connect a 10nF capacitor between BP and SGND to bypass the internal regulator. Do not apply any load to BP. On/Off Control Input. Drive EN high to turn on the MP1567; low to turn it off. For automatic startup, connect EN to IN.
OPERATION
The MP1567 measures the output voltage through an external resistive voltage divider and compares that to the internal 0.9V reference to generate the error voltage at COMP. The current-mode regulator uses the voltage at COMP and compares it to the inductor current to regulate the output voltage. The use of current-mode regulation improves transient response and improves control loop stability. At the beginning of each cycle, the high-side N-Channel MOSFET is turned on, forcing the inductor current to rise. The current at the drain of the high-side MOSFET is internally measured and converted to a voltage by the current sense amplifier. That voltage is compared to the error voltage at COMP. When the inductor current raises sufficiently, the PWM comparator turns off the high-side switch and turns on the low-side switch, forcing the inductor current to decrease. The average inductor current is controlled by the voltage at COMP, which in turn, is controlled by the output voltage. Thus the output voltage controls the inductor current to satisfy the load. Since the high-side N-Channel MOSFET requires voltage above VIN to drive its gate, a bootstrap capacitor from SW to BS is required to drive the high-side MOSFET gate. When SW is driven low (through the low-side MOSFET), the BS capacitor is internally charged. The voltage at BS is applied to the high-side MOSFET gate to turn it on, and maintains that voltage until the high-side MOSFET is turned off and the low-side MOSFET is turned on, and the cycle repeats. Connect a 10nF or greater capacitor from BS to SW to drive the high-side MOSFET gate. Using a larger capacitor does little to improve performance.
MP1567 Rev. 2.3 1/3/2006
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TM
MP1567 - 1.2A SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER
VIN 2.6V to 6V
IN
OFF ON
EN
ENABLE CKT & LDO REGULATOR
GATE DRIVE REGULATOR
Vdr
CURRENT SENSE AMPLIFIER
+ --
BP
VBP 2.4V
PWM COMPARATOR + --
Vdr
BS C7
CONTROL LOGIC
SW Vdr
L1 VOUT
800KHz OSCILLATOR RAMP VBP CURRENT LIMIT COMPARATOR UVLO & THERMAL SHUTDOWN + -+ --
SS C5 -GM -ERROR AMPLIFIER +
PGND
FB
CURRENT LIMIT THRESHOLD SGND COMP R3 C3
VFB 0.9V
MP1567_BD01
Figure 1--Functional Block Diagram
MP1567 Rev. 2.3 1/3/2006
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MP1567 - 1.2A SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER
APPLICATION INFORMATION
COMPONENT SELECTION
Internal Low-Dropout Regulator The internal power to the MP1567 is supplied from the input voltage through an internal 2.4V low-dropout linear regulator, whose output is BP. Bypass BP to SGND with a 10nF or greater capacitor to insure the MP1567 operates properly. The internal regulator cannot supply more current than is required to operate the MP1567, therefore do not apply any external load to BP. Soft-Start The MP1567 includes a soft-start timer that slowly ramps the output voltage at startup to prevent excessive current at the input. This prevents premature termination of the battery voltage at startup due to input current overshoot at startup. When power is applied to the MP1567 a 2A internal current source charges the external capacitor at SS. As the capacitor charges, the voltage at SS will rise. The MP1567 internally limits the feedback threshold voltage at FB to that of the voltage at SS. This forces the output voltage to rise at the same rate as the voltage at SS, forcing the output voltage to ramp linearly from 0V to the desired regulation voltage during soft-start. The soft-start period is determined by the equation:
t SS = 0.45 x C5 R2 = VOUT - VFB VFB R1
Where R2 is the high-side resistor, R1 is the low-side resistor, VOUT is the output voltage and VFB is the feedback regulation threshold. For R1 = 10k and VFB = 0.9V, then R2(k) = 11.1k (VOUT - 0.9V) Selecting the Input Capacitor The input current to the step-down converter is discontinuous, so a capacitor is required to supply the AC current to the step-down converter while maintaining the DC input voltage. A low ESR capacitor is required to keep the noise at the IC to a minimum. Ceramic capacitors are preferred, but tantalum or low ESR electrolytic capacitors will also suffice. Use an input capacitor with a value greater than 10F. The capacitor can be electrolytic, tantalum or ceramic. However, since it absorbs the input switching current it requires an adequate ripple current rating. Use a capacitor with a RMS current rating greater than 1/2 of the DC load current. For insuring stable operation, place the input capacitor as close to the IC as possible. Alternately, a smaller high quality 0.1F ceramic capacitor may be placed closer to the IC with the larger capacitor placed further away. If using this technique, it is recommended that the larger capacitor be a tantalum or electrolytic type. All ceramic capacitors should be placed close to the MP1567. Selecting the Output Capacitor The output capacitor is required to maintain the DC output voltage. Low ESR capacitors are preferred to keep the output voltage ripple to a minimum. The characteristics of the output capacitor also affect the stability of the regulation control system. Ceramic, tantalum or low ESR electrolytic capacitors are recommended. In the case of ceramic capacitors, the impedance at the switching frequency is dominated by the capacitance, and so the
6
Where C5 (in nF) is the soft-start capacitor from SS to GND, and tSS (in ms) is the soft-start period. Determine the capacitor required for a given soft-start period by the equation:
C5 = 2.22 x t SS
Use values for C5 between 10nF and 22nF to set the soft-start period (between 4ms and 10ms). Setting the Output Voltage Set the output voltage by selecting the resistive voltage divider ratio. The voltage divider drops the output voltage to the 0.9V feedback threshold voltage. Use 10k for the low-side resistor of the voltage divider. Determine the high side resistor by the equation:
MP1567 Rev. 2.3 1/3/2006
www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. (c) 2006 MPS. All Rights Reserved.
TM
MP1567 - 1.2A SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER output voltage ripple is mostly independent of the ESR. The output voltage ripple is estimated to be:
f VRIPPLE = 1.4 x VIN x LC f SW
2
Where VRIPPLE is the output ripple voltage, VIN is the input voltage, fLC is the resonant frequency of the LC filter and fSW is the switching frequency. In the case of tantalum or low-ESR electrolytic capacitors, the ESR dominates the impedance at the switching frequency, and so the output ripple is calculated as:
VRIPPLE = I x R ESR
Compensation The system stability is controlled through the COMP pin. COMP is the output of the internal transconductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the characteristics of the control system. The DC loop gain is:
V A VDC = A VEA x G CS x R LOAD x FB V OUT
Where I is the inductor ripple current, and RESR is the equivalent series resistance of the output capacitors. Choose an output capacitor to satisfy the output ripple requirements of the design. A 10F ceramic capacitor is suitable for most applications. Selecting the Inductor The inductor is required to supply constant current to the output load while being driven by the switched input voltage. A larger value inductor results in less ripple current that will results in lower output ripple voltage. However, the larger value inductor has a larger physical size, higher series resistance and/or lower saturation current. Choose an inductor that does not saturate under the worst-case load conditions. A good rule for determining the inductance is to allow the peak-to-peak ripple current to be approximately 30% of the maximum load current. Make sure that the peak inductor current (the load current plus half the peak-to-peak inductor ripple current) is below 2A to prevent loss of regulation due to the current limit. Calculate the required inductance value by the equation:
V x ( VIN - VOUT ) L = OUT VIN x f SW x I
Where AVEA is the transconductance error amplifier voltage gain, GCS is the current sense gain (roughly the output current divided by the voltage at COMP) and RLOAD is the load resistance (VOUT/IOUT where IOUT is the output load current) The system has 2 poles of importance, one is due to the compensation capacitor (C3), and the other is due to the load resistance and the output capacitor (C2). The first is:
fP1 = G EA 2 x A VEA x C3
Where P1 is the first pole and GEA is the error amplifier transconductance (300A/V). The second is:
fP2 = 1 2 x R LOAD x C2
The system has one zero of importance, due to the compensation capacitor (C3) and the compensation resistor (R3). The zero is:
f Z1= 1 2 x R3 x C3
If large value capacitors with relatively high equivalent-series-resistance (ESR) are used, the zero due to the capacitance and ESR of the output capacitor can be compensated by a third pole set by R3 and C4. This pole is:
f P3 = 1 2 x R3 x C4
The system crossover frequency (the frequency where the loop gain drops to 1, or 0dB) is important. Set the crossover frequency to
MP1567 Rev. 2.3 1/3/2006
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MP1567 - 1.2A SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER 75KHz or lower to insure stable operation. Lower crossover frequencies result in slower response and worse transient load recovery. Higher crossover frequencies degrade the phase and/or gain margins and can result in instability. Choosing the Compensation Components The values of the compensation components given in Table 1 yield a stable control loop for the output voltage and capacitor given. Table 1--Compensation Values for Typical Output Voltage/Capacitor Combinations
VOUT C2 R3 C3 C4
Choose the compensation capacitor to set the zero to one fourth of the crossover frequency. Determine the value by the following equation:
C3 = 4 x C2 x VOUT R3 x G EA x G CS x VFB
2
Determine if the second compensation capacitor, C4, is required. It is required if the ESR zero of the output capacitor occurs at less than four times the crossover frequency, or:
8 x C2 x R ESR x f C 1
Where RESR is the equivalent series resistance of the output capacitor. If this is the case, then add the second compensation capacitor. Determine the value by the equation:
C4 = C2 x R ESR(MAX ) R3
1.8V 2.5V 3.3V 1.8V 2.5V 3.3V 1.8V 2.5V 3.3V
4.7F Ceramic 4.7F Ceramic 4.7F Ceramic 10F Ceramic 10F Ceramic 10F Ceramic 47F Tantalum (300m) 47F Tantalum (300m) 47F Tantalum (300m)
3.3k 5.1k 6.8k 7.5k 10k 10k 10k 10k 10k
2.2nF 1.5nF 1.2nF 1nF 820pF 820pF 2.2nF 3.3nF 4.7nF
None None None None None None 1.5nF 1.5nF 1.5nF
Where RESR(MAX) is the maximum ESR of the output capacitor. For Example: Given: VOUT = 1.8V C2 = 10F Ceramic (ESR = 10m max.) Calculate:
R3 4.36 x 10 8 (10F) x (1.8 V ) = 7.85k
To optimize the compensation components for conditions not listed in Table 1, use the following procedure. Choose the compensation resistor to set the desired crossover frequency. Determine the value by the following equation:
R3 = 2 x C2 x VOUT x f C G EA x G CS x VFB
(Use the nearest standard value of 7.5k.)
C3 = 1.9 x 10 -14 = 1.05nF 10F x 1.8 V
(Use 1nF since it is a standard value.)
8 x C2 x R ESR x f C = 0.19
Putting in the known constants and setting the crossover frequency to the desired 75KHz:
R3 4.36 x 10 8 x C2 x VOUT
which is less than 1, therefore the second compensation capacitor (C4) is not required.
In this case, the actual crossover frequency is less than the desired 75KHz, and it is calculated by:
fC = R3 x G EA x G CS x VFB 2 x C2 x VOUT
MP1567 Rev. 2.3 1/3/2006
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MP1567 - 1.2A SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER External Boost Diode For 5V input or output applications, it is recommended that an external boost diode be added. This will help improve the regulator efficiency. The diode can be a low cost diode such as an IN4148 or BAT54.
5V BOOST DIODE 10nF
3
MP1567_F02
BS
1
MP1567
SW
Figure 2--External Boost Diode This diode is also recommended for high duty cycle operation (when
VOUT >65%) and high VIN
output voltage (VOUT>12V) applications.
TYPICAL APPLICATION CIRCUITS
INPUT 6V
2 1 BS SW 3 7 9
C7 10nF
IN EN
OFF ON OPEN IF NOT USED C5 10nF
10 6
MP1567
SS 5 4
FB BP 8
VOUT 1.8V/1.2A
SGND PGND COMP
C3 1nF
C6 10nF C4 OPEN
MP1567_F03
Figure 3--6V Input Application Circuit
MP1567 Rev. 2.3 1/3/2006
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MP1567 - 1.2A SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER
PACKAGE INFORMATION
MSOP10
MP1567 Rev. 2.3 1/3/2006
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TM
MP1567 - 1.2A SYNCHRONOUS RECTIFIED STEP-DOWN CONVERTER
QFN10 (3mm x 3mm)
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications. Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not assume any legal responsibility for any said applications.
MP1567 Rev. 2.3 1/3/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. (c) 2006 MPS. All Rights Reserved.
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