View TPA6203A1_1006081.PDF datasheet online --- IC-ON-LINE

Datasheet File OCR Text:

www.ti.com
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
1.25-W MONO FULLY DIFFERENTIAL AUDIO POWER AMPLIFIER
FEATURES D 1.25 W Into 8 From a 5-V Supply at
THD = 1% (Typ)
APPLICATIONS D Designed for Wireless or Cellular Handsets
and PDAs
D Low Supply Current: 1.7 mA typ D Shutdown Control <1 A D Only Five(1) External Components
- Improved PSRR (90 dB) and Wide Supply Voltage (2.5 V to 5.5 V) for Direct Battery Operation - Fully Differential Design Reduces RF Rectification - Improved CMRR Eliminates Two Input Coupling Capacitors - C(BYPASS) Is Optional Due to Fully Differential Design and High PSRR
DESCRIPTION
The TPA6203A1 is a 1.25-W mono fully differential
amplifier designed to drive a speaker with at least 8- impedance while consuming less than 37 mm2 total printed-circuit board (PCB) area in most applications. This device operates from 2.5 V to 5.5 V, drawing only 1.7 mA of quiescent supply current. The TPA6203A1 is available in the space-saving 2 mm x 2 mm MicroStar Junior BGA package.
Features like 85-dB PSRR from 90 Hz to 5 kHz,
D Avaliable in a 2 mm x 2 mm MicroStar
Junior BGA Package - ZQV (Pb - free) and GQV Options
improved RF-rectification immunity, and small PCB area makes the TPA6203A1 ideal for wireless handsets. A fast start-up time of 4 s with minimal pop makes the TPA6203A1 ideal for PDA applications.
APPLICATION CIRCUIT
RF RI - In From DAC + RI RF SHUTDOWN (1)C(BYPASS) (1) C(BYPASS) is optional B1 C1 Bias Circuitry C3 IN- C2 IN+ VDD A3 Cs _ + VO+ B3 VO- A1 GND B2 RI RI (1)CB
Actual Solution Size
To Battery RF
CS 5,25 mm
RF 6,9 mm
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. MicroStar Junior is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters.
Copyright 2002 - 2003, Texas Instruments Incorporated
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
www.ti.com
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range unless otherwise noted(1) TPA6203A1 Supply voltage, VDD Input voltage, VI Continuous total power dissipation Operating free-air temperature, TA Junction temperature, TJ Storage temperature, Tstg ZQV Lead temperature 1 6 mm (1/16 Inch) from case for 10 seconds 1,6 GQV -0.3 V to 5.5 V -0.3 V to VDD +0.3V See Dissipation Rating Table -40C to 85C -40C to 125C -65C to 150C 260C
235C (1) Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
RECOMMENDED OPERATING CONDITIONS
MIN Supply voltage, VDD High-level input voltage, VIH Low-level input voltage, VIL Common-mode input voltage, VIC Operating free-air temperature, TA SHUTDOWN SHUTDOWN VDD = 2.5 V, 5.5 V, CMRR -60 dB 0.5 -40 2.5 2 0.8 VDD-0.8 85 TYP MAX 5.5 UNIT V V V V C
DISSIPATION RATINGS
PACKAGE GQV, ZQV TA 25C POWER RATING 885 mW DERATING FACTOR 8.8 mW/C TA = 70C POWER RATING 486 mW TA = 85C POWER RATING 354 mW
ORDERING INFORMATION
PACKAGED DEVICES MicroStar Junior (GQV) Device Symbolization TPA6203A1GQVR AADI MicroStar Junior (ZQV) TPA6203A1ZQVR
AAEI (1) The GQV is the standard MicroStar Junior package. The ZQV is a lead-free option and is qualified for 260 lead-free assembly. (2) The GQV and ZQV packages are only available taped and reeled. The suffix R designates taped and reeled parts.
2
www.ti.com
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
ELECTRICAL CHARACTERISTICS
TA = 25C, Gain = 1 V/V PARAMETER |VOO| PSRR CMRR Output offset voltage (measured differentially) Power supply rejection ratio Common-mode Common mode rejection ratio TEST CONDITIONS VI = 0 V, VDD = 2.5 V to 5.5 V VDD = 2.5 V to 5.5 V VDD = 3.6 V to 5.5 V, VIC = 0.5 V to VDD-0.8 VDD = 2.5 V, VIC = 0.5 V to 1.7 V RL = 8 , VIN+ = VDD, VIN- = 0 V or VIN+ = 0 V, VIN- = VDD RL = 8 , VIN+ = VDD, VIN- = 0 V or VIN+ = 0 V, VIN- = VDD VDD = 5.5 V, VI = 5.8 V VDD = 5.5 V, VI = -0.3 V VDD = 2.5 V to 5.5 V, no load, SHUTDOWN = 2 V SHUTDOWN = 0.8 V, VDD = 2.5 V to 5.5 V, no load 1.7 0.01 VDD = 5.5 V VDD = 3.6 V VDD = 2.5 V VDD = 5.5 V VDD = 3.6 V VDD = 2.5 V 4.8 2.1 -90 -70 -62 0.30 0.22 0.19 5.12 3.28 2.24 1.2 1.2 2 0.9 A A mA A V 0.26 MIN TYP MAX 9 -70 -65 -55 0.46 V dB UNIT mV dB
VOL
Low level output Low-level out ut voltage
VOH |IIH| |IIL| IDD IDD(SD)
High level output High-level out ut voltage High-level input current Low-level input current Supply current Supply current in shutdown mode
OPERATING CHARACTERISTICS
TA = 25C, Gain = 1 V/V, RL = 8 PARAMETER TEST CONDITIONS VDD = 5 V VDD = 3.6 V MIN TYP 1.25 0.63 0.3 0.06% 0.07% 0.08% W MAX UNIT
PO
Out ut ower Output power
THD + N 1%, f = 1 kHz N=
VDD = 2.5 V VDD = 5 V, PO = 1 W, f = 1 kHz THD+N Total harmonic distortion plus noise lus VDD = 3.6 V, PO = 0.5 W, f = 1 kHz VDD = 2.5 V, PO = 200 mW, f = 1 kHz C(BYPASS) = 0.47 F, VDD = 3.6 V to 5.5 V, Inputs ac-grounded with CI = 2 F kSVR Supply ripple rejection ratio C(BYPASS) = 0.47 F, VDD = 2.5 V to 3.6 V, Inputs ac-grounded with CI = 2 F f = 217 Hz to 2 kHz, VRIPPLE = 200 mVp-p
-87
f = 217 Hz to 2 kHz, VRIPPLE = 200 mVp-p
-82
dB
SNR Vn CMRR ZI
Signal-to-noise ratio Output lt O t t voltage noise i Common-mode rejection ratio C d j ti ti Input impedance Shutdown attenuation
C(BYPASS) = 0.47 F, f = 40 Hz to 20 kHz, VDD = 2.5 V to 5.5 V, VRIPPLE = 200 mVp-p Inputs ac-grounded with CI = 2 F VDD = 5 V, PO = 1 W No weighting f = 20 H t 20 kH Hz to kHz VDD = 2.5 V to 5.5 V, VICM = 200 mVp-p A weighting f = 20 Hz to 1 kHz f = 20 Hz to 20 kHz
-74
104 17 13 -85 -74 2 -80
dB VRMS V dB M dB
f = 20 Hz to 20 kHz, RF = RI = 20 k
3
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
www.ti.com
MicroStar Juniort (GQV or ZQV) Package (TOP VIEW) GND 12 3 VDD VO+ IN- IN+ (SIDE VIEW)
VO- SHUTDOWN BYPASS
A B C
Terminal Functions
TERMINAL NAME BYPASS GND IN- IN+ SHUTDOWN VDD VO+ VO- NO. C1 B2 C3 C2 B1 A3 B3 A1 I/O I I I I I I O O DESCRIPTION Mid-supply voltage. Connect a capacitor to GND for BYPASS voltage filtering. Bypass capacitor is optional. High-current ground Negative differential input Positive differential input Shutdown terminal. Pull this pin low (0.8 V) to place the device in shutdown and pull it high (2 V) for active mode. Supply voltage terminal Positive BTL output Negative BTL output
4
www.ti.com
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
TYPICAL CHARACTERISTICS TABLE OF GRAPHS
FIGURE vs Supply voltage PO PD Output power Power dissipation Maximum ambient temperature vs Load resistance vs Output power vs Power dissipation vs Output power Total harmonic distortion + noise Supply voltage rejection ratio Supply voltage rejection ratio GSM Power supply rejection GSM Power supply rejection CMRR Common-mode Common mode rejection ratio Closed loop gain/phase Open loop gain/phase IDD Supply current Start-up time vs Frequency vs Common-mode input voltage vs Frequency vs Common-mode input voltage vs Time vs Frequency vs Frequency vs Common-mode input voltage vs Frequency vs Frequency vs Supply voltage vs Shutdown voltage vs Bypass capacitor 1 2, 3 4, 5 6 7, 8 9, 10, 11, 12 13 14, 15, 16, 17 18 19 20 21 22 23 24 25 26 27
5
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
www.ti.com
TYPICAL CHARACTERISTICS
OUTPUT POWER vs SUPPLY VOLTAGE
1.8 1.6 PO - Output Power - W 1.4 1.2 1 0.8 0.6 0.4 0.2 0 2.5 3 3.5 4 4.5 5 THD+N = 1% RL = 8 f = 1 kHz Gain = 1 V/V PO - Output Power - W THD+N = 10% 1.4 1.2 1 0.8 0.6 0.4 0.2 0 8 13 18 23 28 32 RL - Load Resistance - VDD = 5 V VDD = 3.6 V VDD = 2.5 V f = 1 kHz THD+N = 1% Gain = 1 V/V PO - Output Power - W
OUTPUT POWER vs LOAD RESISTANCE
OUTPUT POWER vs LOAD RESISTANCE
1.8 1.6 1.4 VDD = 5 V 1.2 1 0.8 0.6 0.4 0.2 0 8 13 18 23 RL - Load Resistance - 28 32 VDD = 3.6 V VDD = 2.5 V f = 1 kHz THD+N = 10% Gain = 1 V/V
VDD - Supply Voltage - V
Figure 1 POWER DISSIPATION vs OUTPUT POWER
0.4 0.35 PD - Power Dissipation - W 0.3 0.25 0.2 0.15 16 0.1 0.05 0 0 0.2 0.4 0.6 0.8 PO - Output Power - W VDD = 3.6 V 0.6 PD - Power Dissipation - W 8 0.5 0.4 0.3 0.7 VDD = 5 V
Figure 2 POWER DISSIPATION vs OUTPUT POWER
8 Maximum Ambient Temperature - C
Figure 3 MAXIMUM AMBIENT TEMPERATURE vs POWER DISSIPATION
90 80 70 60 50 40 30 20 10 0 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8
16 0.2 0.1 0 0 0.2 0.4 0.6 0.8 1 1.2 1.4 PO - Output Power - W
PD - Power Dissipation - W
Figure 4 TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER
10 5 2 1 0.5 5V 0.2 0.1 0.05 0.02 RL = 8 , f = 1 kHz C(Bypass) = 0 to 1 F Gain = 1 V/V 100 m PO - Output Power - W 1 23 2.5 V 3.6 V
Figure 5 TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER
10 5 2 1 0.5 0.2 0.1 0.05 0.02 0.01 10 m 100 m PO - Output Power - W 1 2 2.5 V 3.6 V 5V RL = 16 f = 1 kHz C(Bypass) = 0 to 1 F Gain = 1 V/V
Figure 6 TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY
THD+N - Total Harmonic Distortion + Noise - % 10 5 2 1 0.5 0.2 0.1 0.05 0.02 0.01 0.005 0.002 0.001 20 1W VDD = 5 V CI = 2 F RL = 8 C(Bypass) = 0 to 1 F Gain = 1 V/V 250 mW 50 mW
THD+N - Total Harmonic Distortion + Noise - %
0.01 10 m
THD+N - Total Harmonic Distortion + Noise - %
100 200 1k 2k f - Frequency - Hz
10 k 20 k
Figure 7 6
Figure 8
Figure 9
www.ti.com
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY
THD+N - Total Harmonic Distortion + Noise - % 10 5 2 1 0.5 0.2 0.1 0.05 0.02 0.01 0.005 0.002 0.001 20 500 mW 125 mW VDD = 3.6 V CI = 2 F RL = 8 C(Bypass) = 0 to 1 F Gain = 1 V/V 25 mW
THD+N - Total Harmonic Distortion + Noise - %
THD+N - Total Harmonic Distortion + Noise - %
TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY
10 5 2 1 0.5 0.2 0.1 0.05 0.02 0.01 0.005 0.002 0.001 20 200 mW 75 mW VDD = 2.5 V CI = 2 F RL = 8 C(Bypass) = 0 to 1 F Gain = 1 V/V 15 mW
TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY
10 5 2 1 0.5 0.2 0.1 0.05 0.02 0.01 0.005 0.002 0.001 20 250 mW VDD = 3.6 V CI = 2 F RL = 16 C(Bypass) = 0 to 1 F Gain = 1 V/V 125 mW
25 mW
50 100 200 500 1 k 2 k f - Frequency - Hz
5 k 10 k 20 k
50 100 200 500 1 k 2 k f - Frequency - Hz
5 k 10 k 20 k
50 100 200 500 1 k 2 k f - Frequency - Hz
5 k 10 k 20 k
Figure 10 TOTAL HARMONIC DISTORTION + NOISE vs COMMON MODE INPUT VOLTAGE
- Supply Voltage Rejection Ratio - dB SVR 10 f = 1 kHz PO = 200 mW
Figure 11 SUPPLY VOLTAGE REJECTION RATIO vs FREQUENCY
-10 -20 -30 -40 -50 -60 -70 -80 -90 -100 20 VDD = 3.6 V 50 100 200 500 1 k 2 k 5 k 10 k 20 k f - Frequency - Hz VDD = 5 V VDD =2. 5 V CI = 2 F RL = 8 C(Bypass) = 0.47 F Vp-p = 200 mV Inputs ac-Grounded Gain = 1 V/V - Supply Voltage Rejection Ratio - dB SVR 0
Figure 12 SUPPLY VOLTAGE REJECTION RATIO vs FREQUENCY
0 -10 -20 -30 -40 -50 -60 -70 -80 -90 -100 20 50 100 200 500 1 k 2 k 5 k 10 k 20 k f - Frequency - Hz VDD = 3.6 V Gain = 5 V/V CI = 2 F RL = 8 C(Bypass) = 0.47 F Vp-p = 200 mV Inputs ac-Grounded VDD =2. 5 V VDD = 5 V
THD+N - Total Harmonic Distortion + Noise - %
1
VDD = 2.5 V 0.10 VDD = 3.6 V
0.01 0 0.5 1 1.5 2 2.5 3 3.5 VIC - Common Mode Input Voltage - V
k
Figure 13 SUPPLY VOLTAGE REJECTION RATIO vs FREQUENCY
- Supply Voltage Rejection Ratio - dB SVR - Supply Voltage Rejection Ratio - dB SVR 0 -10 -20 -30 -40 -50 -60 -70 -80 -90 -100 20 50 100 200 500 1 k 2 k 5 k 10 k 20 k f - Frequency - Hz VDD =2. 5 V VDD = 5 V VDD = 3.6 V CI = 2 F RL = 8 Inputs Floating Gain = 1 V/V
Figure 14 SUPPLY VOLTAGE REJECTION RATIO vs FREQUENCY
- Supply Voltage Rejection Ratio - dB 0 -10 -20 -30 -40 -50 -60 -70 -80 -90 -100 20 50 100 200 500 1 k 2 k 5 k 10 k 20 k f - Frequency - Hz C(Bypass) = 0.47 F C(Bypass) = 0 VDD = 3.6 V CI = 2 F RL = 8 Inputs ac-Grounded Gain = 1 V/V
k
Figure 15 SUPPLY VOLTAGE REJECTION RATIO vs COMMON-MODE INPUT VOLTAGE
-10 -20 -30 -40 -50 -60 -70 -80 -90 0 1 2 3 4 VIC - Common Mode Input Voltage - V 5 VDD = 5 V f = 217 Hz C(Bypass) = 0.47 F RL = 8 Gain = 1 V/V VDD = 2.5 V VDD = 3.6 V
C(Bypass) = 1 F C(Bypass) = 0.1 F
k
Figure 16
k
Figure 17
k
SVR
Figure 18 7
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
www.ti.com
TYPICAL CHARACTERISTICS
0 VDD C1 Frequency 217.41 Hz C1 - Duty 20 % VO - Output Voltage - dBV C1 High 3.598 V C1 Pk-Pk 504 mV VO 0 -50 VDD Shown in Figure 19 CI = 2 F, C(Bypass) = 0.47 F, Inputs ac-Grounded Gain = 1V/V -50
Voltage - V
-100 -150
-100 -150
Ch1 100 mV/div Ch4 10 mV/div t - Time - ms
2 ms/div
0 200 400 600 800 1k 1.2k 1.4k 1.6k 1.8k 2k f - Frequency - Hz
Figure 19 COMMON-MODE REJECTION RATIO vs FREQUENCY
CMRR - Common Mode Rejection Ratio - dB CMRR - Common Mode Rejection Ratio - dB 0 -10 -20 -30 -40 -50 -60 -70 -80 -90 -100 20 50 100 200 500 1 k 2 k f - Frequency - Hz 5 k 10 k 20 k VDD = 2.5 V to 5 V VIC = 200 mVp-p RL = 8 Gain = 1 V/V
Figure 20 COMMON-MODE REJECTION RATIO vs COMMON-MODE INPUT VOLTAGE
0 -10 -20 -30 -40 -50 -60 -70 -80 -90 -100 0 0.5 1 1.5 2 VDD = 3.6 V 2.5 3 3.5 4 4.5 5 VDD = 2.5 V VDD = 5 V RL = 8 Gain = 1 V/V
VIC - Common Mode Input Voltage - V
Figure 21 CLOSED LOOP GAIN/PHASE vs FREQUENCY
40 30 20 10 0 Gain - dB -10 -20 -30 -40 -50 -60 -70 10 100 1k 10 k 100 k 1 M f - Frequency - Hz VDD = 3.6 V RL = 8 Gain = 1 V/V Gain Phase 220 180 140 100 Phase - Degrees 60 20 -20 -60 -100 -140 -180 -220 10 M -100 -150 -200 100 1k 150 100 200
Figure 22 OPEN LOOP GAIN/PHASE vs FREQUENCY
200 VDD = 3.6 V RL = 8 150 100 50 0 -50 Phase Phase - Degrees Gain Gain - dB 50 0 -50
-100 -150 -200 10 M
10 k
100 k
1M
f - Frequency - Hz
Figure 23 8
Figure 24
V DD - Supply Voltage - dBV
GSM POWER SUPPLY REJECTION vs TIME
GSM POWER SUPPLY REJECTION vs FREQUENCY
www.ti.com
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
TYPICAL CHARACTERISTICS
SUPPLY CURRENT vs SUPPLY VOLTAGE
1.8 1.6 I DD - Supply Current - mA I DD - Supply Current - mA 1.4 1.2 1 0.8 0.6 0.4 0.2 0 0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 VDD - Supply Voltage - V 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 1 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2 Voltage on SHUTDOWN Terminal - V VDD = 2.5 V VDD = 3.6 V VDD = 5 V
SUPPLY CURRENT vs SHUTDOWN VOLTAGE
Figure 25 START-UP TIME(1) vs BYPASS CAPACITOR
6 5 Start-Up Time - ms
Figure 26
4
3
2 1 0 0 0.5 1 1.5 C(Bypass) - Bypass Capacitor - F 2
(1) Start-Up time is the time it takes (from a low-to-high transition on SHUTDOWN) for the gain of the amplifier to reach -3 dB of the final gain. Figure 27
9
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
www.ti.com
APPLICATION INFORMATION FULLY DIFFERENTIAL AMPLIFIER
The TPA6203A1 is a fully differential amplifier with differential inputs and outputs. The fully differential amplifier consists of a differential amplifier and a commonmode amplifier. The differential amplifier ensures that the amplifier outputs a differential voltage that is equal to the differential input times the gain. The common-mode feedback ensures that the common-mode voltage at the output is biased around VDD/2 regardless of the commonmode voltage at the input. However, removing the bypass capacitor slightly worsens power supply rejection ratio (kSVR), but a slight decrease of kSVR may be acceptable when an additional component can be eliminated (see Figure 17).
D
Better RF-immunity: GSM handsets save power by turning on and shutting off the RF transmitter at a rate of 217 Hz. The transmitted signal is picked-up on input and output traces. The fully differential amplifier cancels the signal much better than the typical audio amplifier.
Advantages of Fully Differential Amplifiers
D
Input coupling capacitors not required: A fully differential amplifier with good CMRR, like the TPA6203A1, allows the inputs to be biased at voltage other than mid-supply. For example, if a DAC has mid-supply lower than the mid-supply of the TPA6203A1, the common-mode feedback circuit adjusts for that, and the TPA6203A1 outputs are still biased at mid-supply of the TPA6203A1. The inputs of the TPA6203A1 can be biased from 0.5 V to VDD - 0.8 V. If the inputs are biased outside of that range, input coupling capacitors are required. Mid-supply bypass capacitor, C(BYPASS), not required: The fully differential amplifier does not require a bypass capacitor. This is because any shift in the midsupply affects both positive and negative channels equally and cancels at the differential output.
APPLICATION SCHEMATICS
Figure 28 through Figure 30 show application schematics for differential and single-ended inputs. Typical values are shown in Table 1.
Table 1. Typical Component Values
COMPONENT RI RF C(BYPASS)(1) CS CI (1) C(BYPASS) is optional VALUE 10 k 10 k 0.22 F 1 F 0.22 F
D
RF - RI In From DAC + RI RF SHUTDOWN B1 C1 C(BYPASS) Bias Circuitry C3 IN- C2 IN+
VDD A3 Cs _ + VO+ B3 VO- A1 GND B2
To Battery
C(BYPASS) is optional
Figure 28. Typical Differential Input Application Schematic
10
www.ti.com
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
RF CI - IN + CI SHUTDOWN RI RI RF B1 C1 C(BYPASS) Bias Circuitry C3 IN- C2 IN+
VDD A3 Cs _ + VO+ B3 VO- A1 GND B2
To Battery
C(BYPASS) is optional
Figure 29. Differential Input Application Schematic Optimized With Input Capacitors
RF CI IN RI CI SHUTDOWN RF B1 C1 C(BYPASS) Bias Circuitry RI C3 IN- C2 IN+
VDD A3 Cs _ + VO+ B3 VO- A1 GND B2
To Battery
C(BYPASS) is optional
Figure 30. Single-Ended Input Application Schematic Selecting Components Resistors (RF and RI )
The input (RI) and feedback resistors (RF) set the gain of the amplifier according to equation 1. Gain = RF/RI (1)
Bypass Capacitor (CBYPASS ) and Start-Up Time
The internal voltage divider at the BYPASS pin of this device sets a mid-supply voltage for internal references and sets the output common mode voltage to VDD/2. Adding a capacitor to this pin filters any noise into this pin and increases the kSVR. C(BYPASS) also determines the rise time of VO+ and VO- when the device is taken out of shutdown. The larger the capacitor, the slower the rise time. Although the output rise time depends on the bypass capacitor value, the device passes audio 4 s after taken out of shutdown and the gain is slowly ramped up based on C(BYPASS).
RF and RI should range from 1 k to 100 k. Most graphs were taken with RF = RI = 20 k. Resistor matching is very important in fully differential amplifiers. The balance of the output on the reference voltage depends on matched ratios of the resistors. CMRR, PSRR, and the cancellation of the second harmonic distortion diminishes if resistor mismatch occurs. Therefore, it is recommended to use 1% tolerance resistors or better to keep the performance optimized.
Input Capacitor (CI )
The TPA6203A1 does not require input coupling capacitors if using a differential input source that is biased from 0.5 V to VDD - 0.8 V. Use 1% tolerance or better gain-setting resistors if not using input coupling capacitors.
11
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
www.ti.com
In the single-ended input application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the proper dc level. In this case, CI and RI form a high-pass filter with the corner frequency determined in equation 2.
10-F or greater capacitor placed near the audio power amplifier also helps, but is not required in most applications because of the high PSRR of this device.
USING LOW-ESR CAPACITORS
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal) capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance the more the real capacitor behaves like an ideal capacitor.
fc +
1 2p R C II
(2)
-3 dB
DIFFERENTIAL OUTPUT VERSUS SINGLEENDED OUTPUT
Figure 31 shows a Class-AB audio power amplifier (APA) in a fully differential configuration. The TPA6203A1 amplifier has differential outputs driving both ends of the load. There are several potential benefits to this differential drive configuration, but initially consider power to the load. The differential drive to the speaker means that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage swing on the load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power equation, where voltage is squared, yields 4x the output power from the same supply rail and load impedance (see equation 4).
fc
The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit. Consider the example where RI is 10 k and the specification calls for a flat bass response down to 100 Hz. Equation 2 is reconfigured as equation 3.
1 C+ I 2p R f c I
(3)
V V (rms) + V Power +
O(PP) 22 2 (rms) R L
VDD
In this example, CI is 0.16 F, so one would likely choose a value in the range of 0.22 F to 0.47 F. A further consideration for this capacitor is the leakage path from the input source through the input network (RI, CI) and the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high gain applications. For this reason, a ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications, as the dc level there is held at VDD/2, which is likely higher than the source dc level. It is important to confirm the capacitor polarity in the application.
(4)
VO(PP)
Decoupling Capacitor (CS )
The TPA6203A1 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-seriesresistance (ESR) ceramic capacitor, typically 0.1 F to 1 F, placed as close as possible to the device VDD lead works best. For filtering lower frequency noise signals, a
12
RL VDD
2x VO(PP)
-VO(PP)
Figure 31. Differential Output Configuration
www.ti.com
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
In a typical wireless handset operating at 3.6 V, bridging raises the power into an 8- speaker from a singled-ended (SE, ground reference) limit of 200 mW to 800 mW. In sound power that is a 6-dB improvement--which is loudness that can be heard. In addition to increased power there are frequency response concerns. Consider the single-supply SE configuration shown in Figure 32. A coupling capacitor is required to block the dc offset voltage from reaching the load. This capacitor can be quite large (approximately 33 F to 1000 F) so it tends to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting low-frequency performance of the system. This frequency-limiting effect is due to the high pass filter network created with the speaker impedance and the coupling capacitance and is calculated with equation 5.
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased dissipation is understandable considering that the BTL configuration produces 4x the output power of the SE configuration.
FULLY DIFFERENTIAL AMPLIFIER EFFICIENCY AND THERMAL INFORMATION
Class-AB amplifiers are inefficient. The primary cause of these inefficiencies is voltage drop across the output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from VDD. The internal voltage drop multiplied by the average value of the supply current, IDD(avg), determines the internal power dissipation of the amplifier. An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 33).
VO
fc +
1 2p R C LC
(5)
For example, a 68-F capacitor with an 8- speaker would attenuate low frequencies below 293 Hz. The BTL configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency performance is then limited only by the input network and speaker response. Cost and PCB space are also minimized by eliminating the bulky coupling capacitor.
VDD V(LRMS) VO(PP) IDD VO(PP) IDD(avg)
CC RL
-3 dB
Figure 33. Voltage and Current Waveforms for BTL Amplifiers
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different. Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform. The following equations are the basis for calculating amplifier efficiency.
13
fc
Figure 32. Single-Ended Output and Frequency Response
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
www.ti.com
Efficiency of a BTL amplifier + Where:
P P
L
(6)
SUP
2
V V V rms 2 , and V + P , therefore, P + P P+L LRMS L L R 2R 2 L L P sin(t) dt + 1 p R 0 L
pV
1 and P SUP + VDD I DDavg and I DDavg + p Therefore, V DD P SUP pR L substituting PL and PSUP into equation 6, P + Efficiency of a BTL amplifier + Where: V P + 2P R LL VP 2 RL 2 V DD V P p RL
2
V
P R L
[cos(t)] 0 + p RP L
p
2V
2V
+
p VP 4 VDD
PL = Power delivered to load PSUP = Power drawn from power supply VLRMS = RMS voltage on BTL load RL = Load resistance VP = Peak voltage on BTL load IDDavg = Average current drawn from the power supply VDD = Power supply voltage BTL = Efficiency of a BTL amplifier
Therefore, h BTL + p 2P R LL 4V DD
(7)
Table 2. Efficiency and Maximum Ambient Temperature vs Output Power in 5-V 8- BTL Systems
Output Power (W) 0.25 0.50 1.00 1.25 Efficiency (%) 31.4 44.4 62.8 70.2 Internal Dissipation (W) 0.55 0.62 0.59 0.53 Power From Supply (W) 0.75 1.12 1.59 1.78 Max Ambient Temperature (C) 62 54 58 65
Table 2 employs equation 7 to calculate efficiencies for four different output power levels. Note that the efficiency of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at full output power is less than in the half power range. Calculating the efficiency for a specific system is the key to proper power supply design. For a 1.25-W audio system with 8- loads and a 5-V supply, the maximum draw on the power supply is almost 1.8 W. A final point to remember about Class-AB amplifiers is how to manipulate the terms in the efficiency equation to the utmost advantage when possible. Note that in equation 7, VDD is in the denominator. This indicates that as VDD goes down, efficiency goes up.
14
A simple formula for calculating the maximum power dissipated, PDmax, may be used for a differential output application:
P Dmax +
2V2
DD
(8)
p 2R L
PDmax for a 5-V, 8- system is 634 mW. The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor for the 2 mm x 2 mm Microstar Junior package is shown in the dissipation rating table (see page 2). Converting this to JA:
JA
+
(9) 1 1 + + 113C W 0.0088 Derating Factor
www.ti.com
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
Given JA, the maximum allowable junction temperature, and the maximum internal dissipation, the maximum ambient temperature can be calculated with the following equation. The maximum recommended junction temperature for the TPA6203A1 is 125C.
PCB LAYOUT
In making the pad size for the BGA balls, it is recommended that the layout use solder-mask-defined (SMD) land. With this method, the copper pad is made larger than the desired land area, and the opening size is defined by the opening in the solder mask material. The advantages normally associated with this technique include more closely controlled size and better copper adhesion to the laminate. Increased copper also increases the thermal performance of the IC. Better size control is the result of photo imaging the stencils for masks. Small plated vias should be placed near the center ball connecting ball B2 to the ground plane. Added plated vias and ground plane act as a heatsink and increase the thermal performance of the device. Figure 34 shows the appropriate diameters for a 2mm X 2mm MicroStar Junior BGA layout. It is very important to keep the TPA6203A1 external components very close to the TPA6203A1 to limit noise pickup. The TPA6203A1 evaluation module (EVM) layout is shown in the next section as a layout example.
T A Max + T J Max * JA P Dmax + 125 * 113(0.634) + 53.3C
(10)
Equation 10 shows that the maximum ambient temperature is 53.3C at maximum power dissipation with a 5-V supply. Table 2 shows that for most applications no airflow is required to keep junction temperatures in the specified range. The TPA6203A1 is designed with thermal protection that turns the device off when the junction temperature surpasses 150C to prevent damage to the IC. Also, using more resistive than 8- speakers dramatically increases the thermal performance by reducing the output current.
0.38 mm 0.25 mm
0.28 mm
C1
B1
A1
C2
B2
VIAS to Ground Plane
C3
B3
A3
Solder Mask Paste Mask (Stencil) Copper Trace
Figure 34. MicroStar Junior BGA Recommended Layout
15
TPA6203A1
SLOS364B - MARCH 2002 - REVISED JUNE 2003
www.ti.com
TPA6203A1 EVM PCB Layers
The following illustrations depict the TPA6203A1 EVM PCB layers and silkscreen. These drawings are enlarged to better show the routing. Gerber plots can be obtained from any TI sales office.
Only Required Circuitry for Most Applications
Figure 35. TPA6203A1 EVM Top Layer (Not to Scale)
Figure 36. TPA6203A1 EVM Bottom Layer (Not to Scale)
16
MECHANICAL DATA
MPBG144C - JUNE 2000 - REVISED FEBRUARY 2002
GQV (S-PBGA-N8)
PLASTIC BALL GRID ARRAY
2,10 SQ 1,90
1,00 TYP 0,50
C B A 0,50 A1 Corner 1 2 3 1,00 TYP
Bottom View 0,77 0,71
1,00 MAX
Seating Plane 0,35 0,25 0,05 M 0,08
0,25 0,15
4201040/E 01/02 NOTES: A. B. C. D. All linear dimensions are in millimeters. This drawing is subject to change without notice. MicroStar Junior configuration Falls within JEDEC MO-225
MicroStar Junior is a trademark of Texas Instruments.
POST OFFICE BOX 655303
* DALLAS, TEXAS 75265
1
IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services at any time and to discontinue any product or service without notice. Customers should obtain the latest relevant information before placing orders and should verify that such information is current and complete. All products are sold subject to TI's terms and conditions of sale supplied at the time of order acknowledgment. TI warrants performance of its hardware products to the specifications applicable at the time of sale in accordance with TI's standard warranty. Testing and other quality control techniques are used to the extent TI deems necessary to support this warranty. Except where mandated by government requirements, testing of all parameters of each product is not necessarily performed. TI assumes no liability for applications assistance or customer product design. Customers are responsible for their products and applications using TI components. To minimize the risks associated with customer products and applications, customers should provide adequate design and operating safeguards. TI does not warrant or represent that any license, either express or implied, is granted under any TI patent right, copyright, mask work right, or other TI intellectual property right relating to any combination, machine, or process in which TI products or services are used. Information published by TI regarding third-party products or services does not constitute a license from TI to use such products or services or a warranty or endorsement thereof. Use of such information may require a license from a third party under the patents or other intellectual property of the third party, or a license from TI under the patents or other intellectual property of TI. Reproduction of information in TI data books or data sheets is permissible only if reproduction is without alteration and is accompanied by all associated warranties, conditions, limitations, and notices. Reproduction of this information with alteration is an unfair and deceptive business practice. TI is not responsible or liable for such altered documentation. Resale of TI products or services with statements different from or beyond the parameters stated by TI for that product or service voids all express and any implied warranties for the associated TI product or service and is an unfair and deceptive business practice. TI is not responsible or liable for any such statements. Following are URLs where you can obtain information on other Texas Instruments products and application solutions: Products Amplifiers Data Converters DSP Interface Logic Power Mgmt Microcontrollers amplifier.ti.com dataconverter.ti.com dsp.ti.com interface.ti.com logic.ti.com power.ti.com microcontroller.ti.com Applications Audio Automotive Broadband Digital Control Military Optical Networking Security Telephony Video & Imaging Wireless Mailing Address: Texas Instruments Post Office Box 655303 Dallas, Texas 75265 Copyright 2003, Texas Instruments Incorporated www.ti.com/audio www.ti.com/automotive www.ti.com/broadband www.ti.com/digitalcontrol www.ti.com/military www.ti.com/opticalnetwork www.ti.com/security www.ti.com/telephony www.ti.com/video www.ti.com/wireless

▲Up To Search▲

Price & Availability of TPA6203A1

	To Download TPA6203A1 Datasheet File
If you can't view the Datasheet, Please click here to try to view without PDF Reader .