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| (R) ISL6442 Data Sheet April 7, 2006 FN9204.1 Dual (180 Out-of-Phase) PWM and Linear Controller The ISL6442 is a high-performance, triple output controller that provides a single high-frequency power solution primarily for Broadband, DSL and Networking applications. This device integrates complete control, monitoring and protection functions for two synchronous buck PWM controllers and one linear controller. Input voltage ripple and total RMS input current is substantially reduced by synchronized 180 out-of-phase operation of the two PWMs. The two PWM buck converters provide simple voltage mode control. The output voltage of the converters can be precisely regulated to as low as 0.6V, with a maximum tolerance of 1.5% over temperature and line variations. Programmable switching frequency up to 2.5MHz provides fast transient response and small external components. The linear controller provides a low-current output. The ISL6442 has voltage-tracking capability. Each controller has soft-start and independent enable functions combined on a single pin. A capacitor from SS/EN to ground sets the soft-start time; pulling SS/EN pin below 1V disables the controller. Both outputs can soft-start into a pre-biased load. The ISL6442 incorporates robust protection features. An adjustable overcurrent protection circuit monitors the output current by sensing the voltage drop across the upper MOSFET rDS(ON). Hiccup mode overcurrent operation protects the DC/DC converters from damage under overload and short circuit conditions. A PGOOD signal is issued when soft-start is complete and PWM outputs are within 10% of their regulated values and the linear regulator output is higher than 75% of its nominal value. Thermal shut-down circuitry turns the device off if the IC temperature exceeds 150C. Features * 4.5V to 5.5V or 5.5V to 24V Input Voltage Range * 1.5% PWM Switcher Reference Accuracy Over Line and Temperature * Three Programmable Power Output Voltages - Two PWM Controllers with Out-of-Phase Operation - Voltage-Mode PWM Control - One Linear Controller * Programmable Switching Frequency from 300kHz to 2.5MHz * Fast Transient Response - High-Bandwidth Error Amplifier * Extensive Circuit Protection Functions - Overvoltage, Undervoltage, and Overtemperature - Programmable Overcurrent Limit with Hiccup Mode Operation - Lossless Current Sensing (no Sense Resistor needed) * Externally Adjustable Soft-Start Time - Independent Enable Control - Voltage Tracking Capability - Able to Soft-Start into a Pre-Biased Load * PGOOD Output with Delay * 24 Ld QSOP * Pb-Free Plus Anneal Available (RoHS Compliant) Applications * Complete 1 Chip Solution for DSL Modems/Routers * DSP, ASIC, and FPGA Point of Load Regulation * ADSL, Broadband and Networking Applications Pinout ISL6442 (QSOP) TOP VIEW OCSET1 1 SS1/EN1 2 COMP1 3 FB1 4 RT 5 SGND 6 LCDR 7 LCFB 8 FB2 9 COMP2 10 SS2/EN2 11 OCSET2 12 24 VIN 23 BOOT1 22 UGATE1 21 PHASE1 20 LGATE1 19 VCC 18 PGND 17 LGATE2 16 PHASE2 15 UGATE2 14 BOOT2 13 PGOOD Ordering Information PART NUMBER* ISL6442IA ISL6442IAZ (Note) PART TEMP. MARKING RANGE (C) ISL6442IA ISL6442IAZ -40 to 85 -40 to 85 PACKAGE PKG. DWG. # 24 Ld QSOP M24.15 24 Ld QSOP M24.15 (Pb-free) * Add "-TK" to suffix for tape and reel. NOTE: Intersil Pb-free plus anneal products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate termination finish, which are RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. 1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2006. All Rights Reserved All other trademarks mentioned are the property of their respective owners. ISL6442 Block Diagram VCC VIN VCC REFERENCE BIAS CURRENT 30A 100A 0.6V POWER ON RESET AND CONTROL 5V LINEAR REGULATOR 110A OCSET1 VCC UVP1 OVP1 PG1 EN1 COMP1 OUTPUT1 DRIVERS GATE CONTROL LOGIC VCC DEAD-TIME CONTROL OVERCURRENT PGND RAMP1 0 DEG BOOT1 UGATE1 PHASE1 FB1 0.6V FAULT1 UVP1 OVP1 PG1 EN1 SS1 PWM1 LGATE1 VCC5 30A VCC5 30A STARTUP SS1/EN1 SS1 1V DET SS2/EN2 SS2 EN2 EN1 CLOCK AND SAWTOOTH GENERATOR 110A OCSET2 VCC UVP2 OVP2 PG2 EN2 OUTPUT2 DRIVERS PWM2 GATE CONTROL LOGIC VCC DEAD-TIME CONTROL OVERCURRENT BOOT2 FAULT2 UVP2 OVP2 PG2 EN2 SS2 0.6V FB2 RAMP2 180 DEG UGATE2 PHASE2 LGATE2 COMP2 FAULT3 PGND PG3 RT LCFB PGOOD PG1 PG2 PG3 0.6V LCFB LCDR SGND PGND FIGURE 1. BLOCK DIAGRAM 2 FN9204.1 April 7, 2006 ISL6442 Typical Application Schematics ISL6442 VOLTAGE INPUTS REQUIRED VIN (4.5 to 24V) = VIN1 = VIN2 VCC (5V; internal if VIN > 5.6V) VIN3 (<= VCC) for linear TYPE3 COMPENSATION SHOWN C102 VCC COMP1 R102 VOUT1 R101 FB1 UGATE1 VOUT1 PHASE1 LGATE1 Q102 L100 COUT1 C101 OCSET1 BOOT1 COCSET1 CBOOT1 Q101 CIN1 VIN CVCC CVIN VCC VIN OPTIONAL CONNECTION (FOR VIN = VCC = 5V) ROCSET1 VIN1 = VIN R103 C103 C202 R100 COMP2 R202 VOUT2 R201 FB2 C201 ROCSET2 VIN2 = VIN R203 C203 R200 ISL6442 OCSET2 COCSET2 CBOOT2 Q201 UGATE2 CIN2 VOUT2 PHASE2 L200 BOOT2 TYPE3 COMPENSATION SHOWN VCC RPGOOD LGATE2 Q202 PGOOD RT R303 SS1/EN1 SS2/EN2 LCFB LCDR R302 C301 Q301 VIN3 COUT2 CIN3 VOUT3 R301 R300 COUT3 CSS1/EN1 CSS2/EN2 RRT SGND PGND FIGURE 2. TYPICAL APPLICATION 3 FN9204.1 April 7, 2006 ISL6442 Absolute Maximum Ratings (Note 1) SS1/EN1, SS2/EN2, COMP1, COMP2 to SGND . . . -0.3V to +6.0V VCC, FB1, FB2, RT, PGOOD to SGND . . . . . . . . . . . -0.3V to +6.0V LCDR, LCFB to SGND. . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6.0V VIN, OCSET1, and OCSET2 to PGND . . . . . . . . . . . . -0.3V to +28V BOOT1 and BOOT2 to PGND . . . . . . . . . . . . . . . . . . . -0.3V to +33V BOOT1 to PHASE1, and BOOT2 to PHASE2 . . . . . . -0.3V to +6.0V UGATE1 to PHASE1 . . . . . . . . . . . . . . . . . -0.3V to (BOOT1 +0.3V) UGATE2 to PHASE2 . . . . . . . . . . . . . . . . . -0.3V to (BOOT2 +0.3V) LGATE1, LGATE2 to PGND . . . . . . . . . . . . . . -0.3V to (VCC+0.3V) PHASE1, PHASE2 to PGND . . . . . . . . . . . . . . . . . . . . . . -1V to 28V SGND to PGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to 0.3V Thermal Information Thermal Resistance (Typical, Note 2) JA (C/W) QSOP Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . 85 Maximum Junction Temperature (Plastic Package). . -55C to 150C Maximum Storage Temperature Range . . . . . . . . . . . -65C to 150C Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . . 300C Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . . .-40C to +85C Recommended Operating Conditions Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . . .-40C to +85C VCC Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5V 10% VIN Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5V to 24V CAUTION: Stresses above those listed in "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTES: 1. All voltages are measured with respect to GND. 2. JA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details. 3. Guaranteed by design only; not production tested. 4. Design guideline only; not production tested Electrical Specifications PARAMETER VIN SUPPLY Input Operating Supply Current Operating Conditions Unless Otherwise Noted: VIN = 12V, or VCC = 5V 10%, TA = -40C to +85C. Typical values are at 25C. SYMBOL TEST CONDITIONS MIN TYP MAX UNITS ICC_op VIN = 5.5V or 12V; LGATEx, UGATEx Open, FB forced above regulation point (no switching) VIN = 24V - 4.5 7.5 mA - 50 1.25 70 2 mA mA Input Standby Supply Current VCC INTERNAL REGULATOR Output Voltage Maximum Output Current VCC Current Limit (Note 3) REFERENCE AND SOFT-START Reference Voltage at FB1, FB2 ICC_sb VIN = 5.5V, 12V, 24V; SS1/EN1 = SS2/EN2 = 0V VVCC IICC_max IICC_CL VIN > 5.5V VIN = 12V VCC is pulled to PGND; (Note 4) 4.5 80 - 5.2 300 5.5 - V mA mA VREF1, VREF2 VIN = 5V or 12V; TA = 25C VIN = 5V or 12V; TA = 0C to +85C VIN = 5V or 12V; TA = -40C to +85C 0.5925 0.5900 0.5940 0.5915 20 0.8 0.6000 0.6015 30 1.0 50 0.6085 0.6085 0.6100 0.6100 40 1.2 - V V V V V V A V mV Reference Voltage at FB1, FB2 VREF1, VREF2 VIN = 24V; TA = 25C VIN = 24V; TA = 0C to +85C VIN = 24V; TA = -40C to +85C ENx/SSx Soft-Start Current ENx/SSx Enable Threshold ENx/SSxEnable Threshold Hysteresis ISSx VENx VENx_hys (Note 3) - 4 FN9204.1 April 7, 2006 ISL6442 Electrical Specifications PARAMETER ENx/SSx Soft-Start Top of Ramp Voltage POWER-ON RESET ON VCC Rising Threshold Falling Threshold PWM CONVERTERS Minimum UGATE on Time Maximum Duty Cycle Maximum Duty Cycle FBx Pin Bias Current OSCILLATOR Low End Frequency FSW VIN = 5V or 12V; RT = 52.3k VIN = 24V; RT = 52.3k High End Frequency FSW VIN = 5V; RT = 5.23k VIN = 12V; RT = 5.23k Frequency Adjustment Range FSW RT = 52.3k; (Note 3) RT = 5.23k; (Note 3) PWM Sawtooth Ramp Amplitude (peak-peak) VPP (Note 4) 270 270 2.25 2.25 300 305 2.5 2.55 0.3 2.5 1.25 330 340 2.75 2.85 kHz kHz MHz MHz MHz MHz V tUGATE_min (Note 3) DCmax DCmax IFBx VIN = 4.5 or 12V; FSW = 300kHz VIN = 4.5V; FSW = 2.5MHz (Note 3) 95 80 100 80 ns % % nA VPOR_r VPOR_f 4.2 3.85 4.4 4.0 4.475 4.1 V V Operating Conditions Unless Otherwise Noted: VIN = 12V, or VCC = 5V 10%, TA = -40C to +85C. Typical values are at 25C. (Continued) SYMBOL VSSx_top (Note 3) TEST CONDITIONS MIN TYP 3.2 MAX UNITS V PWM CONTROLLER GATE DRIVERS Gate Drive Peak Current Rise Time Fall Time Dead Time Between Drivers ERROR AMPLIFIERS DC Gain Gain-Bandwidth Product Slew Rate Maximum Output Voltage Gain GBWP SR VEA_max (Note 4) (Note 4) (Note 4); COMP = 10pF VCC = 5V; RL = 10k to ground 3.9 88 15 5 4.4 dB MHz V/s (Note 3) (Note 3); CL = 1000pF (Note 3); CL = 1000pF (Note 3) 0.7 20 20 30 A ns ns ns PROTECTION and OUTPUT MONITOR Overvoltage Threshold Undervoltage Threshold OCSET Current Source LINEAR CONTROLLER Drive Sink Current LCFB Feedback Threshold ILCDR VLCFB LCDR TA = 25C TA = -40C to +85C LCFB Input Leakage Current ILCFB (Note 3) 50 0.570 0.595 80 0.620 mA V V nA OV UV IOCSET VOCSET = 4.5V 113 78 80 116 82 110 121 88 140 % % A 5 FN9204.1 April 7, 2006 ISL6442 Electrical Specifications PARAMETER PGOOD Power-Good Lower Threshold Power-Good Higher Threshold Power-Good Lower Threshold PGOOD Delay PGOOD Leakage Current PGOOD Voltage Low THERMAL Shutdown Temperature Shutdown Hysteresis (Note 4) (Note 4) 150 20 C C PG_lowx PG_hix PG_low3 tPGOOD IPGOOD VPG_low LCFB = VCC, LDO disabled PGOOD for Ch1 and Ch2 only LCFB = VCC, LDO disabled PGOOD for Ch1 and Ch2 only LDO enabled, PGOOD for LDO; Ch1 and Ch2 disabled; (Note 3) (Note 3); FSW = 1.4MHz Vpullup = 5.5V IPGOOD = -4mA 88 107 91 110 75 370 94 113 5 0.5 % % % ms A V Operating Conditions Unless Otherwise Noted: VIN = 12V, or VCC = 5V 10%, TA = -40C to +85C. Typical values are at 25C. (Continued) SYMBOL TEST CONDITIONS MIN TYP MAX UNITS Typical Performance Curves 0.65 0.64 REFERENCE VOLTAGE (V) 0.63 0.62 0.61 0.60 0.59 0.58 0.57 0.56 0.55 -40 -20 0 20 40 TEMPERATURE (C) 60 80 0.5 0.0 0 5 10 15 20 RT (k) 25 30 35 40 FSW (MHz) 4.0 3.5 3.0 2.5 2.0 1.5 1.0 FIGURE 3. REFERENCE VOLTAGE VARIATION OVER TEMPERATURE FIGURE 4. FREQUENCY vs RT RESISTOR 6 FN9204.1 April 7, 2006 ISL6442 Pin Descriptions BOOT1, 2 (Pins 23, 14) - These pins power the upper MOSFET drivers of each PWM converter. The anode of the each internal bootstrap diode is connected to the VCC pin. The cathode of the bootstrap diode is connected to this pin, which should also connect to the bootstrap capacitor. UGATE1, 2 (Pins 22, 15) - These pins provide the gate drive for upper MOSFETs, bootstrapped from the VCC pin. PHASE1, 2 (Pins 21, 16) - These are the junction points of the upper MOSFET sources, output filter inductor and lower MOSFET drains. Connect these pins accordingly to the respective converter. LGATE1, 2 (Pins 20, 17) - These are the outputs of the lower N-Channel MOSFET drivers, sourced from the VCC pin. PGND (Pin 18) - This pin provides the power ground connection for the lower gate drivers. This pin should be connected to the source of the lower MOSFET for PWM1 and PWM2 and the negative terminals of the external input capacitors. FB1, 2 (Pins 4, 9) - These pins are connected to the feedback resistor divider and provide the voltage feedback signals for the respective controller. They set the output voltage of the converter. In addition, the PGOOD circuit and OVP circuit use these inputs to monitor the output voltage status. COMP1, 2 (Pins 3, 10) - These pins are the error amplifier outputs for the respective PWM. They are used, along with the FB pins, as the compensation point for the PWM error amplifier. PGOOD (Pin 13) - This is an open drain logic output used to indicate the status of the output voltages. This pin is pulled low when either of the two PWM outputs is not within 10% of the respective nominal voltage or when the linear output drops below 75% of its nominal voltage. To maintain the PGOOD function if the linear output is not used, connect LCFB to VCC. SGND (Pin 6) - This is the signal ground, common to both controllers, and must be routed separately from the high current grounds (PGND). All voltage levels are measured with respect to this pin. VIN (Pin 24) - This pin powers the controllers with an internal linear regulator (if VIN > 5.5V) and must be closely decoupled to ground using a ceramic capacitor as close to the VIN pin as possible. VIN is also the input voltage applied to the upper FET of both converters. TABLE 1. INPUT SUPPLY CONFIGURATION INPUT 5.5V to 24V PIN CONFIGURATION Connect the input supply to the VIN pin. The VCC pin will provide a 5V output from the internal voltage regulator. Connect the input supply to the VCC pin. 5V 10% VCC (Pin 19) - This pin supplies the bias for the regulators, powers the low side gate drivers and external boot circuitry for high side gate drivers. The IC may be powered directly from a single 5V (10%) supply at this pin; when used as a 5V supply input, this pin must be externally connected to VIN. When VIN > 5.5, VCC is the output of the internal 5V linear regulator output. The VCC pin must always be decoupled to power ground with a minimum of 1F ceramic capacitor, placed very close to the pin. RT (Pin 5) - This is the operating frequency adjustment pin. By placing a resistor from this pin to SGND, the oscillator frequency can be programmed from 300kHz to 2.5MHz. SS1/EN1, 2 (Pins 2, 11) - These pins provide enable/disable and soft-start function for their respective controllers. The output is held off when the pin is pulled to the ground. When the chip is enabled, the regulated 30A pull-up current source charges the capacitor connected from the pin to ground. The output voltage of the converter follows the ramping voltage on the SS/EN pin. Note that if either input is held low during power-up, neither channel will start a soft-start ramp until both are released. Once both outputs are running, then either one can be separately disabled and then enabled. But if both are disabled, that requires that both are released before either starts up. See Soft-Start and Voltage Tracking section for more details. LCFB (Pin 8) - This pin is the feedback pin for the linear controller. An external voltage divider network connected to this pin sets the output voltage of the linear controller. If the linear controller is not used, tie this pin to VCC. LCDR (Pin 7) - Open drain output PNP Transistor Driver. LCDR connects to the base of an external PNP pass transistor to form a positive linear regulator. OCSET1, 2 (Pins 1, 12) - These pins are the overcurrent set points for the respective PWM controllers. Connect a resistor (ROCSET) from this pin to the drain of the upper MOSFET. ROCSET, an internal 100A current source, and the upper MOSFET ON resistance rDS(ON) set the converter overcurrent (OC) trip point according to the following equation: I OCSET * R OCSET I OC = -------------------------------------------------r DS ( ON ) ) IOC includes the DC load current, as well as the ripple current. An overcurrent trip initiates hiccup mode. 7 FN9204.1 April 7, 2006 ISL6442 Functional Description Soft-Start and Voltage Tracking After the VCC pin exceeds its rising POR trip point (nominal 4.4V), the chip operation begins. While the voltage on both SS1/EN1 and SS2/EN2 is below 1.0V, the internal switch between SS1/EN1 and SS2/EN2 is turned on so that the voltage across these two pins is the same. If either pin is held low externally, nothing happens until both pins are released. Then both 30A current sources will start charging up both capacitors in parallel. Once the voltage on both of these pins is above 1.0V, this internal switch is turned off and each 30A internal current source charges its corresponding soft-start capacitor connected to its soft-start pin. The charging continues until the voltage across the soft-start capacitor reaches 3.2V. However, the output voltage reaches its regulation value when the soft-start capacitor voltage reaches 1.6V. Figure 5 shows the typical waveforms for SS2/EN2 and VOUT2; SS1/EN1 and VOUT1 are similar. VOUT2 (1V/DIV) VOUT (1V/DIV) GND> 1.6V 1.0V SS1/EN1 (0.5V/DIV) SS2/EN2 (0.5V/DIV) GND> FIGURE 6. VOLTAGE TRACKING The basic timing equation is shown below: dV t = C * ------I where SS2/EN2 (0.5V/DIV) t is the charge time C is the external capacitance 1.6V dV is the voltage charged I is the charging current (nominal 30A) VOUT2 (2V/DIV) 1.0V From 0.0V to 1.0V, C = (C1 + C2 F); dV = 1V; I = (30 + 30A); for a 0.1F capacitor on each pin, t = 3.3ms. This time represents the delay from when the soft-start ramp begins, until the output voltage ramp begins. Then, from 1.0V to 1.6V, the outputs will ramp individually from zero to full-scale. Use the same equation to calculate the time for each ramp; now if V = 0.6V, C = 0.1F, and I = 30A, then t = 2ms. Finally, there is a delay after 1.6V, until the ramp gets to ~3.2V, which signals that the ramp is done; when both ramps are done, the PGOOD delay begins. Figure 7 shows a typical power-up sequence. VIN turns on and begins to ramp up; once VCC passes the rising POR trip point, the linear output is enabled (with no soft-start ramp). The SS1/EN1 pins also start charging (if not held low externally); after a delay for them to reach 1V, VOUT1 and VOUT2 begin to ramp; they are shown in tracking mode. GND> FIGURE 5. SOFT-START The soft-start ramps for each output can be selected independently, but the ISL6442 also has voltage tracking capability. By selecting the soft-start capacitance to be proportional to the output voltage, the output voltage can be tracked. For example, in Figure 6, SS1 cap = 0.18F and SS2 cap = 0.33F, which match the output voltage ratio (1.8V and 3.3V). So the lower VOUT1 ramp will track with the VOUT2 ramp until they both reach 1.8V; VOUT1 then levels off, while VOUT2 continues rising towards 3.3V. 8 FN9204.1 April 7, 2006 ISL6442 VIN (5V/DIV) held low. Once the outputs are running, either output can be disabled and then enabled again, without affecting the other one that's running. But if both SS/EN pins are held low at the same time, then the internal switch will turn on, and both SS/EN pins must be released before they both start to ramp. GND> VOUT2 (1V/DIV) VOUT1 (1V/DIV) VOUT3 (1V/DIV) GND> The linear output does not have a soft-start ramp; however, it may follow the ramp of its input supply, if timed to coincide with its rise, after the VCC rising POR trip. If the input to the linear is from one of the two switcher outputs, then it will share the same ramp rate as the switcher. PGOOD A group of comparators (separate from the protection comparators) monitor the output voltages (via the FB pins) for PGOOD. Each switcher has an lower and upper boundary (nominally around 90% and 110% of the target value) and the linear has a lower boundary (around 75% of the target). Once both switcher output ramps are done, and all 3 outputs are within their expected ranges, the PGOOD will start an internal timer, with the following formula: t PGOOD = 0.5236 ----------------F SW FIGURE 7. OUTPUT VOLTAGES Figure 8 shows pre-biased outputs before soft-start. The solid blue curve shows no pre-bias; the output starts ramping from GND. The magenta dotted line shows the output prebiased to a voltage less than the final output. The FETs don't turn on until the soft-start ramp voltage exceeds the output voltage; then the output starts ramping seamlessly from there. The cyan dotted line shows the output pre-biased above the final output (but below the OVP (Overvoltage Protection)). The FETs will not turn on until the end of the soft-start ramp; then the output will be quickly pulled down to the final value. If the output is pre-biased above the OVP level, the ISL6442 will go into OVP at the end of soft-start, which will keep the FETs off. The output can recover if the voltage goes below the UV (Undervoltage) trip point, at which time a retry will occur. If successful, the output will ramp back up to the normal level. VOUT1 has the same functionality as described above for VOUT2. Each output should react independently of the other, unless they are related by the circuit configuration. SS2/EN2 (0.5V/DIV) where tPGOOD is the delay time (in sec) FSW is the switching frequency (in MHz) Once the time-out is complete, the internal pull-down device will shut off, allowing the open-drain PGOOD output to rise through an external pull-up resistor, to a 5V (or lower) supply, which signals that the "Power is GOOD". Figure 9 shows the three outputs turning on, and the delay for PGOOD. If any of the conditions is subsequently violated, then PGOOD goes low. Once the voltage returns to the normal region, a new delay will start, after which the PGOOD will go high again. The PGOOD delay is inversely proportional to the clock frequency. If the clock is running as slow as 524kHz, the delay will be one second long. There is no way to adjust the PGOOD delay independently of the clock. PGOOD (5V/DIV) GND> VOUT2 OVER-CHARGED VOUT2 (2V/DIV) VOUT2 PRE-BIASED GND> GND> VOUT3 (2V/DIV) VOUT2 (2V/DIV) GND> FIGURE 8. SOFT-START WITH PRE-BIAS NOTE: Neither output cannot be independently disabled during power-up; both SS/EN pins are pulled low internally during POR, and due to the internal switch, neither will start charging if either pin is still VOUT1 (2V/DIV) GND> FIGURE 9. PGOOD DELAY 9 FN9204.1 April 7, 2006 ISL6442 Monotonic Output During soft-start period, the low side MOSFET is disabled to achieve monotonic output voltage when the inductor current is negative. This also allows ramping up into a pre-charged output voltage. Use the following equation to choose the resistor values. RUP is part of the compensation network for the switchers, and should be selected to be compatible; 1k - 5k is a good starting value. Find FB from the Specification Table for the right condition, plug in the desired value for VOUT, and solve for RLOW. FB RUP R LOW = ----------------------------V OUT - FB Switching Frequency The switching frequency of the ISL6442 is determined by the external resistor placed from the RT pin to SGND. See Figure 10 for a graph of Frequency versus RT Resistance. The Specification Table lists a low end value of 52.3k for 300kHz operation (not shown on graph). Running at both high frequency and high VIN voltages is not recommended, due to the increased power dissipation on-chip (mostly from the internal VCC regulator, which supplies gate drivers). The user should check the maximum acceptable IC temperature, based on their particular conditions. 4.0 3.5 3.0 2.5 FSW (MHz) 2.0 1.5 1.0 0.5 0.0 0 5 10 15 20 RT (k) 25 30 35 40 The maximum duty cycle of the ISL6442 approaches 100% at low frequency, but falls off at higher frequency; see the Specification Table. In addition, there is a minimum UGATE pulse width, in order to properly sense overcurrent. The two switchers are 180 out of phase. The linear output voltage is restricted to approximately a 1V-4V range. VIN3 should be equal or less than VCC (in order to be sure that LCDR can turn off the PNP). Note that the linear output is off until the rising POR trips; it does not have a soft-start ramp, and it does NOT shut off, unless VCC goes back below the falling POR trip point. It is suggested that using one of the switcher outputs as the input to the linear allows it to be ramped and enabled/disabled with that switcher. Protection Mechanisms OCP - (Function independent for both PWM). The overcurrent function protects the PWM Converter from a shorted output by using the upper MOSFET's on-resistance, rDS(ON) to monitor the current. This method enhances the converter's efficiency and reduces cost by eliminating a current sensing resistor. The overcurrent function cycles the soft-start function in a hiccup mode to provide fault protection. A resistor connected to the drain of the upper MOSFET and OCSET pin programs the overcurrent trip level. The PHASE node voltage will be compared against the voltage on the OCSET pin, while the upper MOSFET is on. A current (typically 110A) is pulled from the OCSET pin to establish the OCSET voltage. If PHASE is lower than OCSET while the upper MOSFET is on then an overcurrent condition is detected for that clock cycle. The upper gate pulse is immediately terminated, and a counter is incremented. If an overcurrent condition is detected for 32 consecutive clock cycles, and the circuit is not in soft-start, the ISL6442 enters into the soft-start hiccup mode. During hiccup, the external capacitor on the SS/EN pin is discharged, then released and a soft-start cycle is initiated. During soft-start, pulse termination current limiting is enabled, but the 32-cycle hiccup counter is held in reset until soft-start is completed. Figure 12 shows an example of the hiccup mode. As the SS2/EN2 is pulled below the enable trip point, VOUT2 shuts off, and the voltage goes to GND, at which time the output current goes to zero. As SS2/EN2 rises above the enable trip point, the output tries to turn on, the current spikes up, and then is held constant (by the pulse-termination currentlimiting); the output voltage rises, but not up to the desired FIGURE 10. FREQUENCY vs RT RESISTOR Output Regulation Figure 11 shows the generic feedback resistor circuit for any of the three VOUT's; the VOUT is divided down to equal the reference. All three use a 0.6V internal reference (check the Specification Table for the exact reference value at 24V). The RUP is connected to the VOUT; the RLOW to GND; the common point goes to the FB pin. VOUT RUP RLOW 0.6V FB EA COMP FIGURE 11. OUTPUT REGULATION VOUT must be greater than 0.6V and 2 resistors are needed, and their accuracy directly affect the regulator tolerance. R LOW FB = V OUT ----------------------------------R UP + R LOW 10 FN9204.1 April 7, 2006 ISL6442 value. When the SS2/EN2 ramp reaches ~3V, the cycle repeats, and can continue indefinitely. If the short-circuit is removed, the output will ramp up with the next soft-start, and normal operation will resume. VOUT1 and SS1/EN1 will independently function the same way. constraints and the frequency plan of the end equipment. Smaller space requires higher frequency. This allows the output inductor, input capacitor bank, and output capacitor bank to be reduced in size and/or value. The power supply must be designed such that the frequency and its distribution over component tolerance, time and temperature causes minimal interference in RF stages, IF stages, PLL loops, mixers, etc. Inductor Selection The output inductor is selected to meet the output voltage ripple requirements and minimize the converter's response time to the load transient. The inductor value determines the converter's ripple current, and the ripple voltage is a function of the ripple current. The ripple current and voltage are approximated by the following equations, where ESR is the output capacitance ESR value. V IN - V OUT V OUT I = ------------------------------- * --------------F SW * L V IN GND> VOUT2 (2V/DIV) 0A> IOUT2 (2A/DIV) GND> SS2/EN2 (1V/DIV) VOUT = I x ESR FIGURE 12. OVERCURRENT PROTECTION UVP - (Function independent for both PWM). If the voltage on the FB pin falls to 82% (typical) of the reference voltage for 8 consecutive PWM cycles, then the circuit enters into soft-start hiccup mode. This mode is identical to the overcurrent hiccup mode. The UVP comparator is separate from the one sensing for PGOOD, which should have already detected a problem, before the UVP trips. OVP - (Function independent for both PWM). If voltage on FB pin rises to 116% (typical) of the reference voltage, the lower gate driver is turned on continuously (with diode emulation enabled). If SS_DN (internal soft-start done signal) is true (not in soft-start) and the overvoltage condition continues for 32 consecutive PWM cycles, then that output is latched off with the gate drivers three-stated. The capacitor on the SS/EN pin will not be discharged. The switcher will restart when the SS/EN pin is externally driven below 1V, or if power is recycled to the chip, or when the voltage on the FB pin falls to the 82% (typical) undervoltage threshold - after 8 clock cycles the chip will enter soft-start hiccup mode. The hiccup mode is identical to the overcurrent hiccup mode. The OVP comparator is separate from the one sensing for PGOOD, which should have already detected a problem, before the OVP trips. Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance value reduces the converter's response time to a load transient (and usually increases the DCR of the inductor, which decreases the efficiency). Increasing the switching frequency (FSW) for a given inductor also reduces the ripple current and voltage. One of the parameters limiting the converter's response to a load transient is the time required to change the inductor current. Given a sufficiently fast control loop design, the ISL6442 will provide either 0% or 100% duty cycle in response to a load transient. The response time is the time required to slew the inductor current from an initial current value to the transient current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitor. Minimizing the response time can minimize the output capacitance required. The response time to a transient is different for the application of load and the removal of load. The following equations give the approximate response time interval for application and removal of a transient load: L OUT x I TRAN t RISE = -------------------------------------V IN - V OUT L OUT x I TRAN t FALL = -------------------------------------V OUT Application Guidelines PWM Controller Discussion The PWM must be compensated such that it achieves the desired transient performance goals, stability, and DC regulation requirements. The first parameter that needs to be chosen is the switching frequency, FSW. This decision is based on the overall size 11 where: ITRAN is the transient load current step, tRISE is the response time to the application of load, and tFALL is the response time to the removal of load. With a +5V input source, the worst case response time can be either at the application or removal of load and dependent upon the output voltage setting. Be sure to check both of these equations at the minimum and maximum output levels for the worst case response time. Finally, check that the inductor Isat rating is sufficiently above the maximum output current (DC load plus ripple current). FN9204.1 April 7, 2006 ISL6442 Output Capacitor Selection An output capacitor is required to filter the output and supply the load transient current. The filtering requirements are a function of the switching frequency and the ripple current. The load transient requirements are a function of the slew rate (di/dt) and the magnitude of the transient load current. These requirements are generally met with a mix of capacitors and careful layout. Modern microprocessors produce transient load rates above 1A/ns. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (effective series resistance) and voltage rating requirements rather than actual capacitance requirements. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. And keep in mind that not all applications have the same requirements; some may need many ceramic capacitors in parallel; others may need only one. Use only specialized low-ESR capacitors intended for switching-regulator applications for the bulk capacitors. The bulk capacitor's ESR will determine the output ripple voltage and the initial voltage drop after a high slew-rate transient. An aluminum electrolytic capacitor's ESR value is related to the case size with lower ESR available in larger case sizes. However, the equivalent series inductance (ESL) of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Work with your capacitor supplier and measure the capacitor's impedance with frequency to select a suitable component. In most cases, multiple electrolytic capacitors of small case size perform better than a single large case capacitor. Input Capacitor Selection Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use small ceramic capacitors for high frequency decoupling and bulk capacitors to supply the current needed each time Q1 (upper FET) turns on. Place the small ceramic capacitors physically close to the MOSFETs and between the drain of Q1 and the source of Q2 (lower FET). The important parameters for the bulk input capacitor are the voltage rating and the RMS current rating. For reliable operation, select the bulk capacitor with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage and a voltage rating of 1.5 times is a conservative guideline. The RMS current rating requirement for the input capacitor of a buck regulator is approximately 1/2 the DC load current. Switcher MOSFET Selection VIN for the ISL6442 has a wide operating voltage range allowed, so both FETs should have a source-drain breakdown voltage (VDS) above the maximum supply voltage expected; 20V or 30V are typical values available. The ISL6442 gate drivers (UGATEx and LGATEx) were designed to drive single FETs (for up to ~10A of load current) or smaller dual FETs (up to 4A). Both sets of drivers are sourced by the internal VCC regulator (unless VIN = VCC = 5V, in which case the gate driver current comes from the external 5V supply). The maximum current of the regulator (ICC_max) is listed in the Specification Table; this may limit how big the FETs can be. In addition, the power dissipation of the regulator is a major contributor to the overall IC power dissipation (especially as Cin of the FET or VIN or FSW increases). Since VCC is around 5V, that affects the FET selection in two ways. First, the FET gate-source voltage rating (VGS) can be as low as 12V (this rating is usually consistent with the 20V or 30V breakdown chosen above). Second, the FETs must have a low threshold voltage (around 1V), in order to have its rDS(ON) rating at VGS = 4.5V in the 10m-40m range that is typically used for these applications. While some FETs are also rated with gate voltages as low as 2.7V, with typical thresholds under 1V, these can cause application problems. As LGATE shuts off the lower FET, it does not take much ringing in the LGATE signal to turn the lower FET back on, while the Upper FET is starting to turn on, causing some shoot-through current. So avoid FETs with thresholds below 1V. If the power efficiency of the system is important, then other FET parameters are also considered. Efficiency is a measure of power losses from input to output, and it contains two major components: losses in the IC (mostly in the gate drivers) and losses in the FETs. For low duty cycle applications (such as 12V in to 1.5V out), the upper FET is usually chosen for low gate charge, since switching losses are key, while the lower FET is chosen for low rDS(ON), since it is on most of the time. For high duty cycles (such as 5.0V in to 3.3V out), the opposite may be true. Feedback Compensation Equations This section highlights the design consideration for a voltagemode controller requiring external compensation. To address a broad range of applications, a type-3 feedback network is recommended (see Figure 13). 12 FN9204.1 April 7, 2006 ISL6442 C2 C1 COMP FB C3 R1 R3 VOUT R2 gain, given by dMAXVIN /VOSC , and shaped by the output filter, with a double pole break frequency at FLC and a zero at FCE . For the purpose of this analysis, L and D represent the channel inductance and its DCR, while C and E represent the total output capacitance and its equivalent series resistance. 1 F LC = --------------------------2 L C 1 F CE = ----------------------2 C E ISL6442 FIGURE 13. COMPENSATION CONFIGURATION FOR ISL6442 CIRCUIT Figure 14 highlights the voltage-mode control loop for a synchronous-rectified buck converter, applicable to the ISL6442 circuit. The output voltage (VOUT) is regulated to the reference voltage, VREF. The error amplifier output (COMP pin voltage) is compared with the oscillator (OSC) modified sawtooth wave to provide a pulse-width modulated wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter (L and C). The output filter capacitor bank's equivalent series resistance is represented by the series resistor E. The compensation network consists of the error amplifier (internal to the ISL6442) and the external R1-R3, C1-C3 components. The goal of the compensation network is to provide a closed loop transfer function with high 0dB crossing frequency (F0; typically 0.1 to 0.3 of FSW) and adequate phase margin (better than 45). Phase margin is the difference between the closed loop phase at F0dB and 180. The equations that follow relate the compensation network's poles, zeros and gain to the components (R1, R2, R3, C1, C2, and C3) in Figure 14. Use the following guidelines for locating the poles and zeros of the compensation network: 1. Select a value for R1 (1k to 5k, typically). Calculate value for R2 for desired converter bandwidth (F0). If setting the output voltage via an offset resistor connected to the FB pin, Ro in Figure 14, the design procedure can be followed as presented. V OSC R1 F 0 R2 = -------------------------------------------d MAX V IN F LC C2 COMP R2 E/A + C1 R3 C3 FB Ro R1 2. Calculate C1 such that FZ1 is placed at a fraction of the FLC, at 0.1 to 0.75 of FLC (to adjust, change the 0.5 factor to desired number). The higher the quality factor of the output filter and/or the higher the ratio FCE/FLC, the lower the FZ1 frequency (to maximize phase boost at FLC). 1 C1 = ----------------------------------------------2 R2 0.5 F LC VREF 3. Calculate C2 such that FP1 is placed at FCE. OSCILLATOR VIN PWM CIRCUIT VOSC L VOUT C1 C2 = --------------------------------------------------------2 R2 C1 F CE - 1 UGATE HALF-BRIDGE DRIVE D PHASE C E LGATE 4. Calculate R3 such that FZ2 is placed at FLC. Calculate C3 such that FP2 is placed below FSW (typically, 0.5 to 1.0 times FSW). FSW represents the switching frequency. Change the numerical factor to reflect desired placement of this pole. Placement of FP2 lower in frequency helps reduce the gain of the compensation network at high frequency, in turn reducing the HF ripple component at the COMP pin and minimizing resultant duty cycle jitter. R1 R3 = --------------------F SW ----------- - 1 F LC 1 C3 = ------------------------------------------------2 R3 0.7 F SW ISL6442 EXTERNAL CIRCUIT FIGURE 14. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN The modulator transfer function is the small-signal transfer function of VOUT /VCOMP. This function is dominated by a DC 13 It is recommended a mathematical model is used to plot the loop response. Check the loop gain against the error amplifier's open-loop gain. Verify phase margin results and adjust as necessary. The following equations describe the FN9204.1 April 7, 2006 ISL6442 frequency response of the modulator (GMOD), feedback compensation (GFB) and closed-loop response (GCL): d MAX V IN 1 + s(f) E C G MOD ( f ) = ----------------------------- ---------------------------------------------------------------------------------------2 V OSC 1 + s(f) (E + D) C + s (f) L C 1 + s ( f ) R2 C1 G FB ( f ) = ----------------------------------------------------- s ( f ) R1 ( C1 + C2 ) 1 + s ( f ) ( R1 + R3 ) C3 ---------------------------------------------------------------------------------------------------------------------------C1 C2 ( 1 + s ( f ) R3 C3 ) 1 + s ( f ) R2 --------------------- C1 + C2 G CL ( f ) = G MOD ( f ) G FB ( f ) where, s ( f ) = 2 f j phase margin. The mathematical model presented makes a number of approximations and is generally not accurate at frequencies approaching or exceeding half the switching frequency. When designing compensation networks, select target crossover frequencies in the range of 10% to 30% of the switching frequency, FSW. Linear Regulator Compensation Discussion The linear regulator controller controls an external pass element, typically a PNP bipolar junction transistor; see Figure 16 for reference. The error amplifier in the ISL6442 has approximately 72dB (V) of gain. The linear regulator circuit must be compensated such that the gain of the internal error amplifier crosses through 0dB with a slope of 20dB/decade. This allows easily predictable phase response through the 0dB point. The output circuit has a dominant pole determined by the output capacitance and the combination of the sense resistor and the output resistance of the BJT 1 F P1 = -------------------------------------------------2 * R OUT * C OUT COMPENSATION BREAK FREQUENCY EQUATIONS 1 F Z1 = ------------------------------2 R2 C1 1 F Z2 = -------------------------------------------------2 ( R1 + R3 ) C3 1 F P1 = ---------------------------------------------C1 C2 2 R2 --------------------C1 + C2 1 F P2 = ------------------------------2 R3 C3 Figure 15 shows an asymptotic plot of the DC/DC converter's gain vs. frequency. The actual Modulator Gain has a high gain peak dependent on the quality factor (Q) of the output filter, which is not shown. Using the above guidelines should yield a compensation gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 against the capabilities of the error amplifier. The closed loop gain, GCL, is constructed on the loglog graph of Figure 15 by adding the modulator gain, GMOD (in dB), to the feedback compensation gain, GFB (in dB). This is equivalent to multiplying the modulator transfer function and the compensation transfer function and then plotting the resulting gain. MODULATOR GAIN COMPENSATION GAIN CLOSED LOOP GAIN OPEN LOOP E/A GAIN where 1 R OUT = ----------------------------------------------1 1 ----------------------------------- + ---R301 + R302 r o For most pass elements, ro is approximately 100k. It also has a zero determined by the ESR value of the output capacitor and the Capacitance value of the output capacitor: 1 F Z1 = -------------------------------------------------2 * R ESR * C OUT FZ1 FZ2 GAIN FP1 FP2 The compensation network is composed of R300, C300, the internal circuitry of the ISL6442, (also know as hFE in data sheets) of the pass element, and the Miller capacitance of the pass element. The pole is located at 1 F P2 = --------------------------------2 * R X * C X R2 20 log ------- R1 0 where IN 20 log --------------------------------V OSC GCL d MAX V GFB 1 R X = ----------------------------------------------------------------------1 1 1 -------------- + ------------------- + ----------------------R300 1.20k 320 * and C X = C300 + 180pF + C Miller LOG GMOD LOG FLC FCE F0 FREQUENCY If CMiller is unspecified, use 1000pF. The Zero is located at 1 F Z2 = ----------------------------------------------------------2 * ESR C300 * C300 FIGURE 15. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN A stable control loop has a gain crossing with close to a -20dB/decade slope and a phase margin greater than 45. Include worst case component variations when determining 14 FN9204.1 April 7, 2006 ISL6442 Q300 VIN3 CIN3 R300 C300 LCFB R301 R302 COUT3 VOUT3 VIN R307 LCDR Q2. The circuit traces for the MOSFETs' gate and source connections from the ISL6442 must be sized to handle up to 2A peak current. ISL6442 UGATE Q1 LOUT VOUT FIGURE 16. LINEAR COMPENSATION COMPONENTS PHASE LGATE PGND RETURN Q2 Strategy 1. The output capacitor of the linear regulator circuit must be chosen such that the ESR Zero is less than 200kHz F Z1 < 200kHz COUT 2. The voltage divider can be chosen to sink 250A to 1.5mA of sense current, but this is simply a guideline, not a rule. The values should be chosen such that 0.6V R302 = -----------I sen V OUT3 - 0.6V R301 = ------------------------------------I sen FIGURE 17. PRINTED CIRCUIT BOARD POWER AND GROUND PLANES OR ISLANDS CVCC VCC BOOT CBOOT C IN +VIN Q1 L OUT VOUT LOAD ISL6442 SS RT PHASE +VIN VIN SGND PGND RRT CSS SGND CVIN PGND Q2 COUT where Isen = is the current through the resistor divider, and 0.6V is the internal voltage reference that LCFB will equal. 3. Compute the pole and zero for the linear regulator circuit from the Equations above. 4. Make F Z1 F P2 = --------10 FIGURE 18. PRINTED CIRCUIT BOARD POWER AND GROUND PLANES OR ISLANDS 5. Fix R300 at 100. Solve for C300. Use an MLCC, COG or NPO type capacitor for this value. Layout Considerations As in any high frequency switching converter, layout is very important. Switching current from one power device to another can generate voltage transients across the impedances of the interconnecting bond wires and circuit traces. These interconnecting impedances should be minimized by using wide, short printed circuit traces. The critical components should be located as close together as possible using ground plane construction or single point grounding. Figure 17 shows the critical power components of the converter. To minimize the voltage overshoot the interconnecting wires indicated by heavy lines should be part of ground or power plane in a printed circuit board. The components shown in Figure 17 should be located as close together as possible. Please note that the capacitors CIN and COUT each represent numerous physical capacitors. Locate the ISL6442 within 1 inch of the MOSFETs, Q1 and Figure 18 shows the circuit traces that require additional layout consideration. Use single point and ground plane construction for the circuits shown. Locate the RT resistor as close as possible to the RT pin and the SGND pin. Provide local decoupling between VCC and GND pins. For each switcher, minimize any leakage current paths on the SS/EN pin and locate the capacitor, CSS close to the SS/EN pin because the internal current source is only 30A. All of the compensation network components for each switcher should be located near the associated COMP and FB pins. Locate the capacitor, CBOOT as close as practical to the BOOT and PHASE pins (but keep the noisy PHASE plane away from the IC (except for the PHASE pin connection). The OCSET circuits (see Figure 2) should have a separate trace from the upper FET to the OCSET R and C; that will more accurately sense the VIN at the FET than just tying them to the VIN plane. The OCSET R and C should be placed near the IC pins. 15 FN9204.1 April 7, 2006 LOAD CIN ISL6442 Shrink Small Outline Plastic Packages (SSOP) Quarter Size Outline Plastic Packages (QSOP) N INDEX AREA H E -B1 2 3 SEATING PLANE -AD -CA 0.25 0.010 h x 45 L GAUGE PLANE 0.25(0.010) M BM M24.15 24 LEAD SHRINK SMALL OUTLINE PLASTIC PACKAGE (0.150" WIDE BODY) INCHES SYMBOL A A1 A2 B C D A2 C 0.10(0.004) C AM BS MILLIMETERS MIN 1.35 0.10 0.20 0.18 8.55 3.81 5.80 0.26 0.41 24 0 MAX 1.75 0.25 1.54 0.30 0.25 8.74 3.98 6.19 0.49 1.27 8 NOTES 9 3 4 5 6 7 Rev. 2 6/04 MIN 0.053 0.004 0.008 0.007 0.337 0.150 0.228 0.0099 0.016 24 0 MAX 0.069 0.010 0.061 0.012 0.010 0.344 0.157 0.244 0.0196 0.050 8 A1 e B 0.17(0.007) M E e H h L N 0.025 BSC 0.635 BSC NOTES: 1. Symbols are defined in the "MO Series Symbol List" in Section 2.2 of Publication Number 95. 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension "D" does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension "E" does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. "L" is the length of terminal for soldering to a substrate. 7. "N" is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. Dimension "B" does not include dambar protrusion. Allowable dambar protrusion shall be 0.10mm (0.004 inch) total in excess of "B" dimension at maximum material condition. 10. Controlling dimension: INCHES. Converted millimeter dimensions are not necessarily exact. All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation's quality certifications can be viewed at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 16 FN9204.1 April 7, 2006 |
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